09_chapter4SPACEVECTORPULSEWIDTHMODULATION.pdf

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CHAPTER 4
SPACE VECTOR PULSE WIDTH MODULATION
4.1 INTRODUCTION
Multilevel inverters generate sinusoidal voltages from discrete
voltage levels, and pulse width modulation (PWM) strategies accomplish this
task of generating sinusoids of variable voltage and frequency. Modulation
methods for Hybrid Multilevel Inverter can be classified according to the
switching frequency methods. Many different PWM methods have been
developed to achieve the following: Wide linear modulation range, less
switching loss, reduced Total Harmonic Distortion (THD) in the spectrum of
switching waveform: and easy implementation and less computation time.
The most widely used techniques for implementing the pulse with modulation
(PWM) strategy for multilevel inverters are Sinusoidal PWM (SPWM) and
space vector PWM (SPWM). The SVPWM is considered as a better technique
of PWM implementation as it has advantages over SPWM in terms of good
utilization of dc bus voltage, reduced switching frequency and low current
ripple is presented in Beig et al (2007), Gupta and Khambadkone (2007), and
Franquelo et al (2006).
SVPWM is considered a better technique of PWM implementation,
as it provides the following advantages,
(i) Better fundamental output voltage.
(ii) Useful in improving harmonic performance and reducing
THD.

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(iii) Extreme simplicity and its easy and direct hardware
implementation in a Digital Signal Processor (DSP).
(iv) SVPWM can be efficiently executed in a few microseconds,
achieving similar results compared with other PWM methods.
In this chapter, a space vector is defined in a two-dimensional (2-D)
plane and a SVM is performed in the 2-D plane. Furthermore, a three-
dimensional (3-D) space vector has been defined in this chapter for cascaded
H-bridge multilevel inverter. All the existing space vector modulation
schemes are implemented in a two-dimensional, and are therefore unable to
deal with the zero-sequence component caused by unbalanced load.
Complexity and computational cost of traditional SVPWM technique increase
with the number of levels of the inverter as most of the space vector
modulation algorithms proposed in the literature involve trigonometric
function calculations or look-up tables. Previous works on three-dimensional
space vector modulation algorithms have been presented in Prats et al (2003)
and Oscar Lopez et al (2008) for diode-clamped inverter. However, unequal
dc sources cannot be applied to diode-clamped inverter. Meanwhile, the first
3-D space vector modulation for cascaded H-bridge inverter is presented in
Karthikeyan and Chenthur Pandian (2011), which is capable of dealing with
zero-sequence component caused by unbalanced load.
The three-dimensional space vector modulation schemes are
supersets of, and thus are compatible with, conventional two-dimensional
space vector modulation schemes. A new optimized 3-D SVPWM (3-D
OSVPWM) technique was proposed by Karthikeyan and Chenthur Pandian
(2011), which is similar to already existing 3-D SVPWM presented in,
following a similar notation. The proposed SVPWM technique calculate the
nearest switching vectors sequence to the reference vector and the on-state
durations of the respective switching state vectors by means of simple

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addition and comparison operation, without using trigonometric function
calculations, look-up tables or coordinate system transformations. Such very
low complexity and computational cost make them very suitable for
implementation in low cost devices. It is important to notice that these 3-D
OSVPWM techniques can be applied with balanced and unbalanced systems.
Implementation of the 2-D SVPWM and 3-D OSVPWM techniques is carried
out. Both SVPWM algorithms are implemented into a Field Programmable
Gate Arrays (FPGA) from Xilinx Foundation. Matlab Simulink is used to
develop all simulation works. Finally, both algorithmic implementations have
been tested with a cascaded H-bridge multilevel inverter.
4.2 SVPWM FOR THREE-LEG VOLTAGE SOURCE
INVERTER
The topology of a three-leg voltage source inverter is shown in
Figure 4.1. Eight possible switching combinations are generated by the
switching network shown in Figure 4.1 Six out of these eight topologies
producing a nonzero output voltage are known as the non-zero switching
states and the remaining two topologies producing zero output voltage are
known as zero switching states.
Figure 4.1 Three-Phase Voltage Source Inverter

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4.2.1 Voltage Space Vectors
Space Vector Modulation (SVM) for three-leg VSI is based on the
representation of the three phase quantities as vectors in a two-dimensional
( , ) plane. Considering topology 1 of Figure 4.2, which is repeated in
Figure 4.3 a. The line voltages ab
V , bc
V and ca
V are given by
0
-
ab dc
bc
ca dc
V V
V
V V
(4.1)
This can be represented in the ( , ) plane as shown in Figure
4.3(b), where voltages ab
V , bc
V and ca
V are three line voltage vectors
displaced 1200
in space. The effective voltage vector generated by this
topology is represented as 1
V (pnn) in Figure 4.3.b. The switching network
shown in Figure 4.1 has a total of eight possible switching combinations.
Each switching combination is shown in Figure 4.2, and is
represented according to the phase leg connection, where ‘p’ denotes that
phase leg is connected to the positive rail of the DC link, and ‘n’ denotes that
phase leg is connected to the negative rail of the DC link. For example,
switching combination ‘pnn’ represents the condition where the phase A
output terminal a
V is connected to the positive DC rail, and phase B and C
output terminals b
V and c
V are connected to the negative DC rail.

65
Figure 4.2 Eight Switching State Topologies of Three-Phase Inverter
Each switching combination results in a set of three phase voltages
at the AC terminal of the switching network. A reference vector 1
V can be
obtained by transforming the reference three-phase voltage into the
plane, as shown in Figure 4.3.b. A balanced three-phase sinusoidal waveform
is obtained when the reference vector is rotating in the plane.

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Proceeding on similar lines the six non-zero voltage vectors
1 6
V V can be shown to assume the positions shown in Figure 4.4. The tips
of these vectors form a regular hexagon (dotted line shown in Figure 4.4). The
area enclosed by two adjacent vectors, within the hexagon, is chosen as a
sector. Thus there are six sectors numbered 1 to 6 in Figure 4.4.
Figure 4.3.a. Topology V1 (100) Voltage Source Inverter
Figure 4.3.b. Topology Representation of , plane

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The output line voltages generated by this topology in Figure 4.5.a
are given by
0
0
0
ab
cb
ca
V
V
V
(4.2)
Figure 4.4 Non-Zero Voltage Vectors in the , Plane.
The output voltages are represented as vectors which have zero
magnitude and hence are referred to as zero-switching state vectors or zero
voltage vectors. The position at origin in the ( , )plane is as shown in Figure
4.5.b. A total of eight vectors are obtained by transforming the three-phase
voltages into the a b coordinate and the same are called switching state
vectors.
4.2.2 Space Vector Modulation
The desired three phase voltages at the output of the inverter could
be represented by an equivalent vector V rotating in the counter clock wise
direction as shown in Figure 4.6.a. The magnitude of this vector is related to

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the magnitude of the output voltage as shown in Figure 4.6.b and the time this
vector takes to complete one revolution is the same as the fundamental time
period of the output voltage.
Figure 4.5.a. Zero Output Voltage Topologies
Figure 4.5.b Representation of the zero voltage vectors in the , plane
When the desired line-to-line output voltage vector V is in sector 1
as shown in Figure 4.7., vector V could be synthesized by the pulse-width
modulation (PWM) of the two adjacent switching state vectors 1
V (pnn) and
2
V (ppn), the duty cycle of each being d1 and d2, respectively, and the zero
vector (V7 (nnn) / V8 (ppp) ) of duty cycle d0:

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Figure 4.6.a Output voltage vector in the , plane
Figure 4.6.b Output Line Voltage
1 2
1 2
je
g
d V d V V mV e (4.3)
1 2 0 1
d d d (4.4)
where, 0 m 0.850 is the modulation index. This would correspond to a
maximum line-to-line voltage of 1.0Vg, which is 15% more than conventional
sinusoidal PWM is presented in Van Der Broeck et al (1988).

70
All SVPWM schemes and most of the other PWM algorithms, use
the Equations 4.3 and 4.4 for the output voltage synthesis. The modulation
algorithms that use non-adjacent switching state vectors have been shown to
produce higher THD and/or switching losses. But some of them, for e.g.
hysteresis modulation, can be very simple to implement and can provide
faster transient response. The duty cycles 1 2
,
d d and 0
d , are uniquely
determined from Figure 4.7, and the equations of 4.3 and 4.4, the only
difference between PWM schemes that use adjacent vectors is the choice of
the zero vector(s) and the sequence in which the vectors are applied within the
switching cycle.
Figure 4.7 Synthesis of the Required Output Voltage Vector in Sector 1
4.3 SPACE VECTOR MODULATION ALGORITHMS
4.3.1 Two-Dimensional Space Vector
For any balanced three-phase variable, a b c
V V V , where V be the
voltage vector, there is a relationship
0
a b c
V V V (4.5)

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The above equation suggests that the three variables could be
mapped into a vector V on the orthogonal plane, where
V V jV (4.6)
The transformation for this orthogonal co-ordinate mapping,
sometime called 3/2 transformation, is expressed as
2
,
T T
a b c
V V T V V V (4.7)
where T is the transformation matrix and is expressed as
1 1
1
2 2 2
0
3 3 3
2 2
T (4.8)
4.3.2 Three-Dimensional Space Vector
For any balanced three-phase inverter, an assumption is always
made that, 0
a b c
V V V , where V be the voltage vector. The three
variables in a-b-c coordinate abc
V can be mapped into a vector V on the
orthogonal plane. Since each variable maintains the equal phase
difference for each phase, zero–sequence current is automatically nullified as
the reference vectors will be on a plane. When the system becomes
unbalanced, there is no zero–sequence component or triple harmonics because
the reference vectors are on a plane. So the three-dimensional space vector
representation and mapping will be similar to that of 2-D one.

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4.3.3 Two-Dimensional SVPWM for Calculating Switching Angles
A two-dimensional scheme for an n-level 3
n cascaded
multilevel inverter is proposed. In the proposed method, a simple algorithm of
forming switching sequence is applied that leads to minimum change in
voltage. An effective hybrid multilevel inverter must ensure that the Total
Harmonic Distortion (THD) in the voltage output waveform is small enough.
An algorithm is proposed for the cascaded multilevel inverter with equal
voltage steps under the space vector modulation. The algorithm results in the
minimal THD of output voltage of the cascaded multilevel inverter with equal
voltage steps.
A new expression of THD is presented to simplify the derivation.
The output voltage of the hybrid multilevel inverter is 2 1
S with the
SVPWM modulation. 1 2 3
, , ...
p P p pn
E E E E indicates the voltage steps in positive
side and 1 2 3
, , ...
n n n nn
E E E E indicates the voltage steps in negative steps.
1 2 3
, , ... n are the switching angles that indicates the on and off instance of
switches inside the inverter are presented in Liang and Nwankpa (1999),
Sirisukprasert et al (2002) and Zhou and Wang (2002).
4.3.3.1 Mathematical formulation
The algorithm can be expressed from the basis waveform by
applying Fourier series analysis, the amplitude of any odd nth
harmonic can be
expressed as,
1
4
cos
n
n k k
k
V E n
n (4.9)

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where n is an odd harmonic and k is the kth
switching angle. The amplitude of
all even harmonics is zero. The modulation index m is defined as,
1
1
4 n
i
i
V
m
E
(4.10)
Vn is the total harmonic component and V1 is the fundamental
harmonic component.
The voltage THD is defined as
2
2
3,5,7,... 1
n
n
V
THD
V
(4.11)
Now, to find the problem and to implement an algorithm for the
following variable inputs of the inverter 1 2 3
, , ... n
V V V V Modulation index term
m. Output of the algorithm 1 2 3
, , ... n such that THD is minimum. The input
voltages 1 2 3
, , ... n
V V V V are from dc sources.
The input m is determined by a controller in multilevel inverter.
The pulse angles 1 2 3
, , ... n are used by the inverter to control the switches.
It is important to note that minimizing voltage THD is desirable in some
applications. In some high power applications one desires to limit each order
harmonic to certain maximum allowed values. For the three phase system, the
triple order harmonic can be cancelled without help of modulation techniques,
yet it is desired to minimize THD for certain applications is presented in Beig
et al (2004), Naik and Udaya (2005) and Khajehoddin et al (2007).
From modulation index m, determine the value of by evaluating
2
1
1
n
k k
k
m e (4.12)

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where,
1
k
k n
l
i
E
e
E
(4.13)
1
1
/2
/ 2
nk
i k
i
k n
i n
i
E E
E E
(4.14)
The switching angles are determined by,
sin
k k (4.15)
The output voltages of the inverter is,
2
1 k k k k k
k
V E V V V V
(4.16)
where V is unit function. By Fourier series expansion,
1,3,5,...
sin
n
n
V V n
where,
1 1
1,3,5,..
4
cos
n n
V E n
n (4.17)
Modulation index m for the basic output voltage
1 1 1
1
cos
4 4 n
m V E n (4.18)
The THD is expressed as
2
2
3,5,7,... 1
THD n
n
V
V
(4.19)

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4.3.3.2 General structure of the algorithm
The control processing unit calculates the basic parameters to apply
a switching state. The input data to the control processing unit is the reference
space vector. During various iterations, the unit determines the sector number,
triangle number of the subhexagon. The sector number and triangle number
identify the correct switching sequence. The flowchart is given for an n-level
inverter and can be used for any n-levels without change.
The input supply is the amplitude of the voltage steps and
modulation index m, the initial value of o . The flow diagram of the proposed
algorithm to find minimum THD is shown in Figure 4.8. The modulation
index mc is calculated for various iterations. The difference between two
modulation index terms is calculated.
Figure 4.8 Flowchart of the Algorithm

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c
m m (4.20)
where,
– Reference value increases (or) decreases the pulse generation in
the pulse generator.
If the difference between the two modulation index terms is less
than reference value , the proposed algorithm outputs the optimal switching
angles. The iteration method is used to solve and find minimization of the
voltage THD.
4.3.3.3 Principles of the 2-D SVPWM technique
The SVPWM technique can be easily extended to all multilevel
inverters by Rodriguez et al (2002). This section explains the 2-D technique
for the generation of SVPWM for a five-level inverter. By using the space-
vector diagram, the basic principles of the SVPWM method can be easily
explained. Figure 4.9 shows the space vector diagram of a five level inverter.
The SVPWM implementation involves two phases: i) Selecting the switching
vector, and ii) Determining the center of the subhexagon.
A. Selection of the Switching Vector
In the 2-D method, the small triangles formed by the adjacent
voltage space vectors are called sectors. Such six sectors around a space
vector forms a hexagon called subhexagon. The space vector modulation
diagram of a multilevel inverter can be viewed composed of a number of
subhexagons. Sector identification is done by determining the triangle that
encloses the tip of the reference space vector.

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Figure 4.9 Space Vector Diagram for a Five Level Inverter
Numerous publications on two-dimensional space vector
modulation strategies are developed and applied to multilevel inverters with
equal dc sources. Previous works from authors on 2-D SVPWM algorithms
presented in Aneesh Mohamed et al (2009) and McGrath et al (2003) shows
that the reference voltage vector is identified from a single sub hexagon. In
the proposed 2-D SVPWM technique, the 2-D algorithm chooses two
subhexagons randomly. From the two chosen subhexagons, the 2-D
algorithm compares and selects the subhexagon that contains the nearest
reference vector.
The selected subhexagon that contains the tip of the reference space
vector is mapped to the inner subhexagon by subtracting the vector located at
the center of the subhexagon, from which switching sequence generations are
carried out. This comparison process helps in determining the appropriate
subhexagon having the nearest reference vector as well as the sector to which
the reference vector is pointing to is determined for calculating the switching
states and the switching times.

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After identification of the sector in which the tip of the reference
vector is located, four adjacent triangles having four nearest state vectors (two
redundant vectors) to the reference vector are identified. The adjacent
switching vectors and its corresponding switching sequence for the multilevel
inverter are determined. Then the algorithm determines the time duration of
each switching sequence. Once the duration of the switching sequence is
calculated, then its corresponding switching angles are determined. The 2-D
algorithm proposed reduces the total harmonic distortion of output voltage of
the inverter as the exact voltage vector is chosen for switching sequence
generation. The proposed modulation is computational very efficient and cost
effective which can be applied to multilevel inverters with any number of
levels.
The shaded region in Figure 4.10 shows two subhexagons. They are
represented as “subhexagon I” having vector 000 as the centre and
“subhexagon II” having the vector 032 as the centre. Another “subhexagon
III” is also considered, having a vector 330 as the centre. The inner
subhexagon can be viewed as a space vector diagram of a two-level inverter
whose inverter voltage vectors switch between the lower most levels.
Subhexagon II can be also viewed as a space vector diagram of a two level
inverter whose voltage vectors involve higher levels.
The shifting of subhexagon in the space vector diagram of a
multilevel inverter to the zero vector 000 simplifies the switching time
calculations associated with multilevel inverters. The shifting of subhexagon
II in the space vector diagram of a multilevel inverter toward the zero vector
000 involves the mapping of the sectors of subhexagon II to the sectors of the
inner subhexagon. This is done by subtracting the vector at the centre of the
subhexagon II from its other vector.

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Figure 4.10 Mapping of Reference Space Vector for Switching Vector
Generation
In a reverse approach of mapping, the inner subhexagon can be
mapped to subhexagon II by adding the voltage space vector at the centre of
subhexagon II to the vector of the inner subhexagon. Consider the voltage
vectors 000, 001, 101 and 111 associated with sector 5 of the inner
subhexagon and the voltage space vector 032 which is the vector at the centre
of subhexagon II. Adding the voltage space vector 032 to the voltage space
vector associated with sector 5 of the inner subhexagon gives the vectors 032
(001+033), 022(101+022) and 421(100+021), which are the vectors
associated with sector 5 of subhexagon.
Also, the voltage space vector associated with any subhexagon can
be generated by adding the vector at the centre of the particular subhexagon to
the voltage space vector of the corresponding sectors in the inner subhexagon.
The mapping of the inner subhexagon to any other outer subhexagon called as
reverse mapping is used to generate the vectors associated with any sector in
the space vector diagram of the multilevel inverter is presented in Aneesh
Mohamed et al (2009).

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B. Determining the Centre of the subhexagon
The space vector diagram of a five-level inverter, shown in Figure
4.11 can be viewed as the form of five levels with four layers. These levels
are represented as Level 1 to 5. The instantaneous reference space vector
lying in layer4 (P=4) and within the S1 region forms a triangular shape.
Depending upon the layer of operation of the instantaneous
reference space vector, all vectors for the center of the subhexagon are
generated, and the vector which is closest to the reference space vector is
taken as the center of the subhexagon. Figure 4.11 also shows the six 60o
regions 1 2 3 4 5
, , , ,
S S S S S and 6
S .
Figure 4.11 Levels in the Space Vector Diagram of a Five-Level Inverter
The subhexagon associated with the instantaneous reference space
vector can be considered as centered on the inner side of layer 4. The
instantaneous reference space vector can be resolved in to the axes ,
x y
V V and
z
V using the following where ,
a b
V V and c
V are the instantaneous amplitude of
the three reference phase voltages

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3
2
3
2
3
2
x a c
y b a
z c b
V V V
V V V
V V V
(4.21)
The axis lying in the 60 region which contains the instantaneous
reference space vector will have maximum magnitude among the values.
4.3.4 3-D Space Vector Pulse Width Modulation
A simple three-dimensional (3-D) space vector algorithm of
multilevel inverters for mitigating harmonic content in three phase systems
was proposed by Prats et al (2003), and Franquelo et al (2006). The three-
dimensional SVPWM (3-D SVPWM) scheme computes the switching state
vectors and the nearest switching sequence. The 3-D modulation technique
allows directly compensating harmonics in three phase systems and
optimizing the switching sequence minimizing the number of switchings.
Most of the SVM algorithms found in the literature for multilevel
inverters use a representation of voltage vectors in coordinates, instead
of using abc coordinates The representation offers an interesting
information about the zero-sequence component of both currents and voltages
(proportional to the gamma coordinate), however the change of reference
frame have to be carried out, implies complex calculations. In addition, the
three-dimensional (3-D) representation of the switching vectors, in is
difficult to understand. Most methods based on representation need to
determine the “sextant” in which the desired voltage vector is included, which
leads to many complicated operations, including rotations, complex
comparisons etc.

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Using coordinates, the possible tetrahedrons that compose the
state vectors space have different shapes and volumes. It is not easy to
develop computationally efficient modulation algorithms to find out the
tetrahedron where the reference vector is pointing to. However, Zhang et al
(2002) has developed 3-D SVPWM algorithms using coordinates for
three-leg four-wire (3L4W) topologies. But these algorithms are complex and
their computational cost is very high.
This is the fundamental drawback of this type of 3-D SVPWM
algorithms. Therefore, it is necessary to change the way of representation for
the multilevel state vectors space. This is the reason because abc coordinates
are used by other authors doing modulation algorithms very simple and more
easily implemented. In order to reduce the 3-D SVPWM computational cost,
multilevel inverters state vectors space can be represented using
abc coordinates instead of coordinates.
The 3-D SVPWM techniques carry out a search of the four nearest
state vectors to determine the switching sequence. Using abc coordinates, the
3-D SVPWM algorithm control complexity and the computational load is
lower than using coordinates. Using abc coordinates, the control region
is a cube or a prism for three-legged inverter. The 3-D SVPWM technique
presented in need to synthesis 3-D vector space and identifies the subcube
where the reference vector is located. Once this subcube is determined, it is
divided into six tetrahedrons, and the 3-D SVPWM has to calculate the
tetrahedron where the reference vector is pointing to.
Finally, the switching sequence and the corresponding duty cycles
are determined. This technique can be used as the modulation algorithm in all
applications needing a 3-D vector control such as an active filter, where the
conventional 2-D SVM cannot be used. The 3-D space vector modulation

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schemes are supersets of, and as a result 3-D SVPWM is compatible with
conventional two dimensional space vector modulation schemes.
4.3.4.1 Synthesis of Reference Vector
The 3-D space vector modulation is to synthesize the reference
vector ref
V using the switching vectors in the abc coordinate. It is similar to
that of 2-D space vector modulation (describe in section 4.2). Synthesis of the
reference vector in abc coordinates needs to take the following steps: (1)
selection of switching vectors and (2) sequencing the switching vectors.
4.3.4.1.1. Selection of Switching Vectors
For the 2-D space vector modulation it is easy to identify the
adjacent switching vectors. For 3-D space vector, it takes the following steps
to identify the adjacent vectors.
1. The 3-D SVPWM techniques carry out a search of the four
nearest state vectors to determine the switching sequence.
Using abc coordinates, the control region is a cube or a prism
for a three-legged inverter. Then the control region is divided
into several sub-cubes. The four vectors nearest to the
reference vector must be identified. The 3-D SVPWM
algorithm easily calculates the four state vectors which
generate the reference vector. The multilevel control region is
a cube, which is divided into several sub-cubes and the first
step of the modulation algorithm is to find the sub-cube where
the reference vector is pointing to. The cubes in three-
dimensional space are formed by a certain number of sub-
cubes depending on the number of the levels of the inverter.
Only one subcube for two-level inverters, eight sub-cubes for

84
three-level inverters, twenty-seven sub-cubes for four-level
inverters. In general, 3
( 1)
n sub-cubes into each cube,
where n is the number of levels of the multilevel inverter.
2. Once this subcube is determined, it is divided into six
tetrahedrons. Thus, the reference vector will be pointing to a
volume which is a tetrahedron. Within each sub-cube, six
tetrahedrons can be identified. The adjacent switching vectors
are defined by the tetrahedrons. Therefore, it is necessary to
define the tetrahedron where the reference vector is pointing
to. This tetrahedron is easily found using comparisons with
three 450
planes into the 3-D space, which define the six
tetrahedrons inside the subcube. The three planes define the
six tetrahedrons. Notice that only a maximum of three
comparisons are needed regardless the inverter number of levels.
3. Once abc coordinates are known, the algorithm calculates the
four state vectors corresponding to the four vertices of the
tetrahedron into the selected sub-cube, which form the
switching sequence. These vectors will generate the reference
vector. Configurations of the 3-D space with different
numbers of tetrahedrons in the cube have been studied.
However, the minimum number of comparisons is obtained
using the six tetrahedrons.
4.3.4.1.2. Sequencing of the Switching Vectors
The SVPWM algorithm calculates the four state vectors into the 3-
D space and the corresponding duty-cycles and a maximum of three
comparisons for calculating the suitable tetrahedron. The computational load
is always the same and it is independent of the number of levels. In addition,

85
the algorithm provides the switching sequence that minimizes the total
harmonic distortion.
4.3.4.2 Calculation of the duty cycles
Once the state vectors which generate each reference vector are
known, the corresponding duty-cycles are calculated. The numeric evaluation
of the duty cycles or on-state durations of the switching states are reduced to a
simple addition. The algorithm generates a matrix S with four state vectors
and the corresponding switching times i
t . Where , ,
i i i
a b c
S S S with 1,..,4
i are
the coordinates of each state vector and i
d is the corresponding duty cycle
1 1 1
1
2 2 2
2
3 3 3
3
4 4 4 4
1,..,4
a b c
a b c
i i m
a b c
a b c
S S S d
S S S d
S t d T i
d
S S S
d
S S S
(4.22)
Where, m
T is the sample time.
The coordinates abc represent the different voltage levels between
each phase and the neutral. They take values between zero and2( 1)
n , where
n is the number of levels of the multilevel inverter. The duty cycles are only
functions of the reference vector components and the integer part of reference
vector coordinates. In addition, the optimized switching sequence is selected
in order to minimize the number of switching.
4.3.5 Optimized Three-Dimensional SVPWM for Balanced and
Unbalanced Systems
A simple and generalized 3-D OSVPWM (optimized 3-D space
vector modulation) is presented for multilevel inverters, which can be used for

86
any number of levels of multilevel inverter. The 3-D OSVPWM scheme
proposed is similar to the existing 3-D SVPWM presented in Prats et al
(2003), and Leon et al (2009), following the similar notation. Since the
proposed scheme is similar to the already existing 3-D SVPWM scheme, once
the duty cycle, which is the function of reference vector components is
calculated. Then it translates the generating vectors into switching pulses. In
general, the SVPWM techniques are used to generate an average voltage
vector equal to the reference voltage vector.
In two-dimensional SVPWM (2-D SVPWM), the space vectors are
contained in a plane when the system is balanced without triple harmonics. A
perfect balance of the dc voltages of a multilevel inverter cannot be achieved
in all loading conditions. Load imbalances or transient loads have a
significant impact on the multilevel inverter dc voltage ripple. In this case, 2-
D SVPWM modulation techniques are not prepared for this unbalance
because they do not take it into account to carry out the modulation process.
As a result, errors appear in the output modulated voltages because they do
not match to the desired output voltages when they are brought to average
over a period of switching.
This fact leads to an increase of the harmonic distortion of the
output voltages and currents of the multilevel inverters. This problem has
been previously addressed by Blaabjerg et al (1999) and Enjeti and Shireen
(1992), avoiding the influence of the dc-link voltage ripple on the output
signals for two-level inverters. Previous works from Parts et al (2003) and
Leon et al (2009) focused on 3-D SVPWM algorithm shows the reference
vectors are not on a plane, if the system is unbalanced for multilevel inverter.
A new three-dimensional space vector modulation scheme named as 3-D
OSVPWM is presented. The simple and generalized 3-D OSVPWM scheme
proposed for multilevel inverter is an improved version of the previous 3-D

87
SVPWM technique because the proposed technique can be used for any
number of levels of the multilevel inverter. The 3-D OSVPWM technique
presented, takes into account the actual unbalance of the multilevel inverter to
carry out the necessary calculations, avoiding errors in the modulation
process. The proposed 3-D OSVPWM algorithm maintains 1200
phase for
each phase to compute switching state vectors and the nearest switching
sequence. The reference vectors are on a plane with 1200
phase shift for each
phase.
Each switching state of the inverter is represented by a switching
voltage vector. The method is to choose the vectors that, applied during a
certain time over the switching period, produce a voltage vector equal to the
reference or desired voltage vector. The proposed 3-D OSVPWM algorithm
permits the on-line selection of the nearest space vectors sequence for
generating the reference voltage vector. The 3-D OSVPWM algorithm readily
computes the nearest switching vectors sequence to the reference vector and
calculates the on-state durations of the respective switching state vectors.
Since each vector maintains an equal phase difference for each phase, the
flow of zero–sequence current is automatically nullified as there will be no
component in the voltage vector and the reference vectors will be contained
on a plane.
When the system becomes unbalanced, there is no zero–sequence
component of both current and voltage (proportional to the coordinate) or
triple harmonics because the reference vectors are on an equal 1200
phase shift
plane. So the 3-D space vector representation and mapping will be similar to
that of 2-D one. This technique can be used as modulation algorithm in all
applications needing a 3-D control vector such as active filters, where the
conventional two dimensional space vector modulations cannot be used. The
unequal dc voltage does not affect the THD of the output waveform, leading

88
to an economical cost and reduction of the harmonic content. The 3-D
OSVPWM can be applied to multilevel inverters with any number of levels.
Using the proposed 3-D OSVPWM, balanced and unbalanced systems can be
modulated with balanced or unbalanced dc voltages.
4.3.5.1 Determination of 3-D Control Region
In the 3-D SVPWM technique explained in Section 4.3.4., the dc
voltages of a multilevel inverter are not balanced. Therefore, the existing 3-D
SVPWM technique cannot be used because the 3-D control region changes
and it is not formed by regular cubes. The 3-D control region of a multilevel
inverter changes because the distinct locations of the switching state vectors
move due unbalance in the voltage. Three different dc voltages 1 2
,
dc dc
V V and
3
dc
V have to be considered for a three-bridge fourteen-level cascaded
multilevel inverter. The various possible Vphase-0 voltages of the multilevel
inverter are 1 1 2
0, ,
dc dc dc
V V V and 1 2 3
dc dc dc
V V V . The phase states can be
represented using generalized dc voltages 1 2
,
dc dc
V V and 3
dc
V .
In general, the 3-D control region formed will be cube with
size 1 2 3
dc dc dc
V V V , which forms several rectangular sub cubes with different
sizes, depending on the voltages of different dc sources. The 3-D control
region of a fourteen-level power inverter having 1 2 3
dc dc dc
V V V is
represented in Figure 4.12. The i values are the size of the sub cubes that
form the 3-D control region, are as follows:
dctotal dc1 dc2 dcn
V =V +V +.....+V
(4.23)
di
i
dctotal
V
=
V (4.24)

89
The actual output voltages of the multilevel inverter are determined
from the vector Vo. The elements of the vector Vo are in increasing order from
zero to the positive value. In the N-level inverter, the vector is represented as
o dc1 dc1 dc2 dc1 dcn-1
V= 0,V ,V +V ,....,V +...+V
(4.25)
Figure 4.12 Three-Dimensional Control Region of a Five –Level Inverter
with Voltage Unbalance in the DC Link Vdc1<Vdc2<Vdc3
Normalizing the vector Vo with respect to the total dc voltage
generating vector Von

90
o
o n
d cto tal
V
V =
V (4.26)
dc1 dc1 dc2 dc1 dcn-1
on
dctotal dctotal dctotal
V V +V V +...+V
V = 0, , ,....,
V V V
(4.27)
1 1 2 1 N -1
= 0, ,....., +...+
(4.28)
1 1 2
= 0 , , . .. . ., 1
(4.29)
4.3.5.2 Reference vector normalization
The reference vector calculated is defined as , ,
refn a b c
V V V V where
j
V is the voltage of phase j with respect to point zero. The reference vector Vref
is normalized using the total dc voltage of the multilevel inverter. The
normalized reference voltages ,
a b
V V and c
V take values between zero and one.
The reference vector in the nth
component is represented as
a c
b
a c
refn b
dctotal dctotal dctotal
V
V V
V ={v , v , v }= , ,
V V V (4.30)
4.3.5.3 Algorithm for generating switching angles
1) Read the instantaneous magnitudes of phase voltages.
2) Determine the coordinates of the instantaneous space vector.
3) The coordinates of the reference space vector is normalized
through division by the normalization constant VDC / n – 1.
4) Determine the tetrahedron region enclosing the normalized
cube for an n-level inverter.

91
5) Calculate the modulation index m, which is determined by a
number of iterations repeatedly applied to various
tetrahedrons.
6) Identify the tetrahedron region having m, calculated in step (5)
7) Determine the centroid of each of the four triangles. Also
determine the tetrahedron with centroid closest to the 16
triangles.
8) Each triangle gives three vectors and switching states.
9) Determine the nearest switching angle and the resultant m
from step (5), go to next step, else go to step (7).
10) The tetrahedron finally determined represents the tetrahedron
enclosing the space vector.
11) Continue the process of identifying the voltage space vector in
different tetrahedron of the cube.
12) Select the zero vector from the vectors located at the vertices
of the identified subcube.
4.3.5.4 Determination of the tetrahedron
Several iterations are carried out over each component to find out
the nearest centroid of the tetrahedral region enclosing the normalized
subcube. Several iterations are carried out in phase a, to find where a
v is
located inside Von vector, comparing with each element. Finally, the lower
and upper closer elements in vector Von of the range where va is located can
be determined. For example, for the four level stages, it is as follows
dc1 dc1 dc2 dc1 dc2 dc3
on
dc1 dc2 dc3 dc1 dc2 dc3 dc1 dc2 dc3
V V +V V +V +V
V = 0, , ,
V +V +V V +V +V V +V +V (4.31)

92
1 1 2 1 2 3
= 0, , ,
(4.32)
1 2
= 0, , , 1
3 3 (4.33)
If 1 < a
v < 1, the factor a
v is 1 and the factor a
Vs is one. For each
phase of the reference vector this process is repeated to calculate the vector
, ,
abc a b c
V V V V which is the nearest centroid of the triangular region where
the reference vector , ,
a b c
V V V is located. Also, the
vector , ,
abc a b c
Vs Vs Vs Vs is determined. Vectors , ,
a b c and
, ,
a b c
V can be calculated.
a b c a a b b c c
={ , , }={Vs -V ,Vs -V ,Vs -V} (4.34)
a b c a a b b c c
V={ , , }={v -V , v -V , v -V} (4.35)
The switching sequences and the duty cycles calculations are
summarized in Table 4.1 and the parameters ,
a b and c in the table are
defined as
a b c
a b a
a b c
= , = , = (4.36)

93
Figure 4.13 Division of Each Rectangular Subcube of the 3-D SVPWM
Control Region of a Multilevel Inverter with Unbalanced
DC Voltages
The proposed modulation technique finds out the four nearest state
vectors to form the switching sequences to generate the reference voltage. The
switching sequence consisting of four nearest state vectors have four
tetrahedron in each sub cube and six different cases is shown in Figure 4.13.
Determine the centroid of each of the four triangles; each triangle gives three
vectors and a switching state.

94
Table 4.1 Switching sequence and Duty Cycles Determined by the 3-D
SVPWM Technique
S. No Cube Cases
State Vector
Sequences
Duty Cycles
1 Case 1
210, 310, 321
210, 310, 320
210, 321, 320
321, 310, 320
D = 1-µa
D = µa - µc
D = µc - µb
D = µb
2 Case 2
210, 212, 220
210, 212, 321
210, 220, 321
212, 321, 220
D = 1-µc
D = µc - µa
D = µa - µb
D = µb
3 Case 3
210, 311, 310
210, 311, 321
210, 310, 321
310, 311, 321
D = 1-µc
D = µc - µb
D = µb - µa
D = µa
4 Case 4
210, 321, 220
210, 320, 220
210, 321, 320
220, 320, 321
D = 1-µb
D = µb - µa
D = µc - µa
D = µa
5 Case 5
210, 211, 212
210, 321, 212
210, 321, 211
212, 321, 211
D = 1-µb
D = µb - µa
D = µa - µc
D = µc
6 Case 6
210, 311, 321
210, 311, 211
210, 211, 321
211, 311, 321
D = 1-µa
D = µa - µc
D = µb - µc
D = µc

95
4.4 2-D SVPWM ALGORITHM COMPONENTS
Figure 4.14 shows the block diagram designed for the 2-D
algorithm, which included the following components: frequency, 2 ,
dq 2-
DSVM, 2 ,
abc PWM3 and dead time.
4.4.1 Component Frequency-
The user can select one of four modulator switching frequencies
available by means of the input signal s
f . The selected frequency defines the
needed values for the switching period T and the dead time Td. A dead time of
5% of the switching period was used.
4.4.2 Component dq to
This element carries out the following to
dq transformation of
the reference vector ,
d q
r r
v v from the rotational frame to the stationary
frame.
-1 cos -sin
3
2 sin cos
0
d
r r
q
r r
v v
v v
(4.37)
4.4.2.1 Component 2-DSVM
This component is the core of the system. It determines the nearest
three vector 1 1 2 2
, , ,
T T
s s s s
v v v v and 3 3
,
T
s s
v v to the reference vector
in the ,
T
r r
v v in the frame and calculates their corresponding
switching times which is presented in Celanovic and Boroyevic (2001).

96
4.4.2.2 Component to abc
This component transforms back three nearest vectors from to
the abcframe. This is achieved by evaluating the Equation 4.37. This
expression has multiple solutions. Therefore, the following algorithm was
developed to obtain a vector sequence that minimizes the number of
switchings. The twenty four triangular regions have been joined in the six
highlighted groups shown in Figure 4.15. The vector sequence starts with the
boxed vector of the group, and it is tailored with adjacent vectors.
The 1 1 1
, ,
T
a b c
s s s
v v v is the boxed vector and vector 2 2 2
, ,
T
a b c
s s s
v v v and
3 3 3
, ,
T
a b c
s s s
v v v are taken accordance with the arrow inside the region.
Figure 4.14 2-DSVPWM Algorithm

97
Figure 4.15 Switching Vector Sequence Selection
The following components PWM3 generates a symmetrical
sequence that starts and ends with the same vector, therefore, there are no
additional switchings when the reference vector changes between triangular
regions that belong to the same group, but it only adds two extra switchings
when the reference changes to the next group. Hence, this vector selection
method minimize the number of switchings when the reference vector stays
inside a region as well as when it changes region.
4.4.2.3 Components PWM3
It arranges the three vectors of the sequence in the symmetrical way
into the switching period. The sequence generated is the following.

98
1
1 1 1
2
2 2 2
3
3 3 3
1 1 1 1
1 1 1
3 3 3
, , - - - - - -
4
, , - - - - - -
2
, , - - - - - -
2
, , - - - - - -
2
, ,
T
a b c
s s s
T
a b c
s s s
T
a b c
s s s
T
a b c
s s s
T
a b c
s s s
T
v v v
T
v v v
T
V V V
T
v v v
V V V 3
2
2 2 2
1
1 1 1
- - - - - -
2
, , - - - - - -
2
, , - - - - - -
4
T
a b c
s s s
T
a b c
s s s
T
T
v v v
T
v v v
(4.38)
After that, it generates the six PWM signals corresponding to each
complementary pair of switches.
4.4.2.4 Components dead-time
Finally, this circuit generates the trigger signal for each power
switch, introducing the proper dead time to give each Metal Oxide
Semiconductor Field Effect Transistor (MOSFET) enough time to switch off,
before its complementary one is switched on. This is done, as shown in Figure
4.17 by delaying the rising edges of the trigger signals by the time Td.
4.4.3 3D SVPWM Components
Figure 4.16 shows the block diagram designed for the 3-D
algorithm, which includes the following components: frequency, 2
dq abc, 3D
SVPWM, PWM4 and dead time.

99
4.4.4 Component Frequency
This block is similar to the one used in the 2D SVPWM algorithm
for calculating the switching period T and the dead time Td.
4.4.5 Component dq to abc
This component implements the following 2
dq abcco-ordinate
transformation
1 0 0
cos -sin
1 3
- 0
sin cos
2 2
1 3
- - 0
2 2
a
r d
r
b
r q
r
c
r
v
v
v
v
V
(4.39)
4.4.5.1 Component 3DPWM
This component finds the four nearest switching vectors
1 1 1 2 2 2 3 3 3 4 4 4
, , , , , , , , , and , ,
T T T T
a b c a b c a b c a b c
s s s s s s s s s s s s
v v v v v v v v v v v v to
the reference vector and calculates their corresponding switching times, and
the resulting sequences minimizes of switches.

100
Figure 4.16 3D SVPWM Algorithm
4.4.5.2 Components of PWM
This block is very similar to the PWM3 but working with four input
vectors instead of three. It arranges the vectors in a symmetrical way into the
switching time and generates the Six PWM signals corresponding to each
complementary pair of MOSFETs. The sequence generated is the following.
1
1 1 1
2
2 2 2
3
3 3 3
4 4 4 4
3
3 3 3
2 2 2
, , - - - - - -
2
, , - - - - - -
2
, , - - - - - -
2
, , - - - - - - -
, , - - - - - -
2
, , - - -
T
a b c
s s s
T
a b c
s s s
T
a b c
s s s
T
a b c
s s s
T
a b c
s s s
T
a b c
s s s
T
v v v
T
v v v
T
v v v
v v v T
T
v v v
v v v 2
1
1 1 1
- - -
2
, , - - - - - -
2
T
a b c
s s s
T
T
v v v
(4.40)

101
4.4.5.3 Component dead time
This circuit generates the trigger signal for each power switch,
introducing the proper dead time to give each MOSFET enough time to
switch off, before its complementary one is switched on.
Figure 4.17 Dead Time Implementation
This is done, as shown in Figure 4.17 and to generate the twelve
trigger signals inserting the corresponding dead times.
4.5 COMPONENT IMPLEMENTATION
The hardware description of the components of the algorithms
shown in Figure 4.9 and 4.11 has been hand-coded in VHDL. An exception
was made with the components dq to anddqtoabc. These components
have been developed in the Simulink with the System Generator libraries
provided by Xilinx. The VHDL code that describes these components has
been automatically generated using the System Generator tool, in order to
produce a correct system implementation in a short design time.
4.5.1 Component Frequency
One of the four frequencies and dead time pairs can be selected by
means of the two-bit signal fs. Signals T and Td are integer numbers that
express the switching period and dead time in microseconds. Signal T is
twelve bits wide, which allows periods of upto 1023µs i.e., 977.5Hz. Td is a

102
six bit signal, which allows defining a dead time upto 63µs. The modulator
switching period T and the corresponding dead time Td are selected by means
of a four-channel multiplexer, as shown in the Figure 4.18.
Figure 4.18 Component Frequency
4.5.2 Components dq to
The implementation of transformation from the rotating to the
stationary frames is given in Equation 4.37. Figure 4.19 shows the Simulink
models used to describe the components and to generate the corresponding
VHDL description files. Internal arithmetic operations are done with adequate
precision using fixed-point number representation and two’s complement
format. The fractional part of the final result was rounded to 8-bits in order to
calculate the switching times with adequate accuracy in the component 2-D
SVPWM. For an n-level inverter in the 2-D SVPWM algorithm, the integer
part of the result takes values from ( 1) ( 1)
n to n . Therefore 3 bit are

103
necessary to represent the integer part. The output signals in the 2-D case are
11-bit wide.
Figure 4.19 Component of dq to
4.5.3 Components dqto abc
The component implements transformation Equation 4.39 from the
rotating to the stationary frames. Figure 4.20 shows the Simulink models used
to describe the components and to generate the corresponding VHDL
description files. Internal arithmetic operations are done with adequate
precision using fixed-point number representation and two’s complement
format. The fractional part of the final result was rounded to 8-bits in order to
calculate the switching times with adequate accuracy in the component 3-D
SVPWM. For an n-level inverter, in the 3-D SVPWM algorithm the integer
parts of the results take values in the range0 ( 1)
to n . Therefore, 2 bits are
necessary to represent the integer part. Consequently, the output signals in the
3-D case are 10 bit wide. The 3-D SVPWM can be used without any
modification for a CHB inverter of upto 5 levels. If the number of level
increases then more bits would be needed to represent these signals properly.
The sine and cosine operations have been implemented by means of a table

104
stored in a memory. 256 points of the sinusoidal waveform have been stored
with 8-bits of resolution. In order to store the data a 256*8 memory is needed.
Therefore an external RAM was used to take advantage of the FPGA
hardware resources.
Figure 4.20 Component dqto abc
4.5.4 Component 2-D SVM
The 2-D SVPWM has been implemented strictly following the
algorithm descriptions given in Celanovic and Boroyevic (2001) by means of
simple arithmetic and comparison operations.
Figure 4.21 Flow Diagram of VHDL Description of the 2-D SVPWM
Component

105
The nearest vector and switching times are calculated from the
integer and fractional part of the reference. The signals corresponding to the
vector component has been 3-bit in the 2-D algorithm and the unit switching
times are 8-bit wide.
The integer and fractional part of the references can be done by the
proper bit extraction from the fixed point number. Corrections have been done
in the case of negative numbers in the 2-D algorithm. Figure 4.21 show the
flow diagram of the VHDL description and the bit extraction operation.
4.5.5 Component 3D SVPWM
The 3-D SVPWM has been implemented strictly following the
algorithm descriptions given in Prats et al (2003) by means of simple
arithmetic and comparison operations.
Figure 4.22 Flow Diagram of VHDL Description of the 3D SVPWM
Component

106
The nearest vector and switching times are calculated from the
integer and fractional part of the reference. The signals corresponding to the
vector component has 2-bit in the 3-D algorithm and the unit switching times
are 8-bit wide. Figure 4.22 shows the flow diagram of the VHDL description
and the bit extraction operation.
4.5.6 Component toabc
Figure 4.23 Flow Diagram of VHDL Description of the to
abccomponent
Figure 4.23 shows the flow diagram utilized for the VHDL
description of this block in accordance with Figure 4.15. As shown
previously, the output signals of the component have 2-bits which are enough
to represent the three levels of inverter.
4.5.7 Components PWM3 and PWM4
Figure 4.19 shows the PWM4 implementation. The implementation
of component PWM3 is very similar. But, in this case unit time T4 is not
needed and the four vector is internally generated as

107
4 4 4 1 1 1
, , 1, 1, 1
T T
a b c a b c
s s s s s s
v v v v v v (4.41)
The block sequence compares the switching time corresponding to
each vector with the value of a counter to generate the vector index
corresponding to each time interval. The 50 MHz master clock of the S3 was
divided using the digital clock manager of the FPGA in order to obtain a 10
KHz clock. That clock allows generating the PWM signals with a time
precision of 0.1 µs. Due to the fact that vector times are 8-bit wide, then the
maximum switching frequency available in the system is 100MHz without the
frequency division of the master clock, the maximum. Switching frequency
could be increased 5 times, upto 125 KHz. The output of the multiplexer is the
space vector that must be generated by the inverter at a particular time.
Figure 4.24 Component Diagram of the PWM4
The level of each phase in the corresponding trigger signals is
translated by the component Level to trigger signal. This translation depends
on the inverter topology, therefore this block must be redesigned if an inverter

108
different from the CHB is used. It is not necessary to modify the rest of
components because both modulation algorithms do not depend on the
topology.
4.5.8 Component Dead-Time
Figure 4.25 Component of Dead Time Description
This block generates the inverse of trigger signals adding the
corresponding dead time. It was implemented using an edge detector together
with a counter working as a timer in order to delay rising edges of the trigger
signal as show in Figure 4.25.
4.6 RESOURCES
All circuits were combined in a top file according to Figure 4.14.
and 4.16. Finally, the whole system is synthesized and implemented in the
XS200E FPGA through the use of the Xilinx foundation ISE tools, which are
specific for these tasks. Both algorithm implementations use only one ERAM,
to generate the sine and cosine functions, but the 2-D algorithm uses half flip-
flops and two hardware multipliers less than 3-D algorithm. This is because
the first algorithm works with only three 2-D vectors, instead of four 3-D
vectors of the second algorithm. Although the 2-D algorithm works with
fewer vectors represented with fewer bits, both implementations use similar
number of logic blocks and look-up tables (LUTs). This is due to the need for

109
an extra task to select the generating vectors in the 2-D algorithm.
Consequently, the 3-D algorithm is easier to implement but uses more logical
resources of the FPGA.
In order to extend the proposed implementation to a higher level
inverter, many considerations have to be taken into account. The
implementation of the component frequency is independent of the inverter
topology; therefore it can be used without any modification. Components
dq to , dq to abc , 2-D SVPWM and 3-D SVPWM only need slight
modifications in order to adapt the bit number of signals to the number of
level of the inverter. Resources used by those components increases with the
increase in the number of levels. In fact, they could be directly used for a five
level topology. Components PWM3, PWM4 and dead-time have to be
extended in order to generate additional PWM signals for extra MOSFET’s of
the inverter. Therefore, resources used by the additional components increases
almost linearly with number of levels of the CHB inverter. The component
to abcwas designed for a five-level inverter.
4.7 COMPARISON OF SVPWM AND PWM
The SVPWM is considered a better technique of PWM implementation
as it has some advantages over SPWM in terms of good utilization of dc-bus
voltage, reduced switching frequency and low current ripple. SVPWM
provides the following advantages: i) better fundamental output voltage; ii)
useful in improving harmonic performance and reducing THD; and iii) easier
hardware implementation in digital signal processor. SVPWM can be
efficiently executed in a few micro seconds, achieving similar results
compared with other PWM methods.

110
The carrier-based PWM methods were developed first and were
widely used in most applications. One of the earliest modulation signals for
carrier-based PWM is sinusoidal PWM (SPWM). The SPWM technique is
based on the comparison of a carrier signal and a pure sinusoidal modulation
signal. The SVPWM technique calculates and computes the duty cycles but
SPWM technique derives through comparison. The utilization rate of the DC
voltage for traditional sinusoidal PWM is only 78.5% of the DC bus voltage,
which is far less than that of the six-step wave (100%). This technique can be
used in single-phase and three-phase inverters. The SVPWM technique can
increase the fundamental component by up to 27.39% that of SPWM. The
fundamental voltage can be increased up to a square wave mode where a
modulation index of unity is reached. The SVPWM is significantly better than
SPWM by approximately 15.5%.
4.8 SUMMARY
In this chapter, two very efficient SVPWM techniques were
discussed. The basic idea about SVPWM for three-legged voltage source
inverter was discussed in detail. It is illustrated how the voltage space vectors
are defined in a two-dimensional (2-D) plane and three-dimensional (3-D)
plane for a cascaded H-bridge multilevel inverter. Furthermore, two-
dimensional (2-D) and three-dimensional (3-D) space vector modulation
algorithms are explained. Mathematical formulation for calculating switching
angles and switching sequence was determined and the Fourier series theory
was used to derive the harmonic equations corresponding to the multilevel
switching scheme.
The basic idea about the existing three-dimensional space vector
modulation schemes was discussed. Furthermore, the optimized three-
dimensional (3-D) space vector modulation for balanced and unbalanced
system was explained. A new 3-D OSVPWM technique is proposed to carry

111
out the necessary calculations taking the actual values of the dc voltages in
the modulation process and therefore undesirable harmonic distortions in
output waveform is avoided in unbalanced condition. The optimized 3-D
space vector modulation algorithm for cascaded H-bridge inverter was
proposed to handle the zero-sequence component caused by unbalanced load.
Using the proposed 3-D OSVPWM, balanced and unbalanced systems can be
modulated with balanced or unbalanced dc voltages. The computational cost
of the proposed modulation technique is very low and the technique can be
applied to multilevel inverter with any number of levels.

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