FIVE-TRANSISTOR SINGLE-PORT SRAM BIT CELL WITH HIGH SPEED AND LOW STANDBY CURRENT

International Journal of VLSI design & Communication Systems (VLSICS) Vol.9, No.4, August 2018
DOI: 10.5121/vlsic.2018.9401 1
FIVE-TRANSISTOR SINGLE-PORT SRAM BIT CELL
WITH HIGH SPEED AND LOW STANDBY CURRENT
Chien-Cheng Yu1,3*
, Ming-Chuen Shiau2
, and Ching-Chih Tsai3
1
Department of Electronic Engineering, Hsiuping University of Science and Technology,
Taichung City, Taiwan
2
Department of Electrical Engineering, Hsiuping University of Science and Technology,
3
Department of Electrical Engineering, National Chung Hsing University
ABSTRACT
In this paper, a new five-transistor (5T) single-port Static Random Access Memory (SRAM) cell with
voltage assist is proposed. Amongst them, a word line suppression circuit is designed to provide a voltage
of the respective connected word line signal in a selected row cells lower than the power supply voltage
VDD by a threshold voltage during a read operation, thereby to improve the read/write-ability of the cell. In
addition, a voltage control circuit is coupled to the sources corresponding to driver transistors of each row
memory cells. This configuration is aimed to control the source voltages of driver transistors under
different operating modes. Specifically, during a read operation, a two-stage reading mechanism is
engaged to increase the reading speed. Simulation results for the proposed cell design confirm that there is
a conspicuous improvement in reading speed and power saving over the conventional SRAM cells, and fast
writing also can be achieved.
KEYWORDS
Single-port, Static random access memory, Assist circuitry, Voltage control circuit, Standby start-up circuit
1. INTRODUCTION
Semiconductor memories can be characterized as volatile random access memories (RAMs) or
non-volatile read only memories (ROMs), where RAMs can either be static (SRAM) or dynamic
(DRAM) differing mainly in the manner by which they store a state of a bit. In applications of
large-scale semiconductor integrated circuits (LSIs), static random access memories (SRAMs) are
the widely-used on-chip memories. Typically, SRAM circuits may be single-port or multi-port. In
the single-port SRAM, normally, either of read and write operation is performed in one access
from one port circuit connected to a pair of bit lines to one memory cell. SRAM is arranged as a
matrix of thousands of individual memory cells fabricated in an integrated circuit chip, and
address decoding in the chip allows access to each cell for read/write functions. In SRAMs, each
memory cell includes transistor-based circuitry that implements a bi-stable latch, which can only
assume one of two possible states, namely on (state 1) or off (state 2). The latch can only be
induced to change from one state to the other through the application of a voltage or other
external stimuli. However, for stabilization of an operation of the conventional six-transistor (6T)
SRAM, the current driving capability ratio between the driver and access transistors should be
maintained at 2 to 3 or more, and the driver transistor should be designed to have a large gate
width, which also causes an increase in size of the memory cell of the SRAM. As such, higher
integration and larger capacity cannot be expected with a conventional 6T SRAM. Furthermore, it
is apparent that the conventional 6T SRAM suffers from the disadvantage of relying on too many

2
transistors. Accordingly, there is an important need to have an SRAM cell that requires fewer
than six transistors.
An SRAM cell has three modes of operation, namely read, write and standby [1]. SRAM cells use
a write operation to store data in the cell and a read operation to sense the data stored in the cell.
In order to perform a write operation, a signal is asserted on word line WL. That is, the voltage on
word line WL will go high. This activates the access transistors MA1 and MA2. Under this state,
one of the bit lines BL and BLB is driven to the logic low level while the other to the logic high
level depending on the data (logical ‘0’ or logical ‘1’) to be written. As a result, the data is written
to the cross-coupled inverters. Furthermore, to perform a read operation, the voltage level of the
word line WL is raised to the logic high level, activating the transistors MA1 and MA2. This
results in producing a voltage level difference between the bit lines BL and BLB depending on
the data held in the cross-coupled inverters. This voltage level difference is amplified by a sense
amplifier (not shown) so as to read the data. At the end of the read and write operations, the word
line voltage is de-asserted to ground allowing the cross-coupled inverters to function normally
and hold the logic state of the storage nodes.
In recent years, there has been an increasingly growing trend towards portable devices, which
increases a demand for lower power consumption of a large-scale semiconductor integrated
circuit (LSI). Leakage current from a memory cell can cause unnecessary power consumption,
especially during a standby mode [2-3]. As CMOS technology scales down to 65 nm and below,
the power consumption caused by leakage currents is becoming a significant part of the global
power consumption [4-5]. Recent researches have shown that the leakage current will become
even greater than the dynamic current in the overall power consumption [6-8]. Typically, there
are three major sources of leakage in a MOS transistor, namely subthreshold leakage, gate
leakage, and reverse bias junction leakage [9-10]. Amongst them, Gate-Induced drain leakage
(GIDL) is an unwanted short-channel effect that occurs at higher drain biases in an overdriven off
state of a MOS transistor. The GIDL is the result of a deep depletion region that forms in the
drain at high drain-to-gate biases. However, Drain-induced barrier lowering (DIBL) is a short-
channel effect in MOS transistors referring originally to a reduction of threshold voltage of the
transistor at higher drain voltages [11]. With scaling down of the MOS transistor, each of the
leakage sources may increase accordingly, thus resulting in the increase of the total leakage
current. Therefore, it would clearly be desirable to provide a design for an SRAM cell that has
less leakage current than conventional designs when the cell in standby.
The remainder of this paper is organized as follows. Section 2 presents a brief description of
conventional 6T and 5T SRAM cell topologies. The proposed 5T SRAM cell with integrated
read/write assist is described in Section 3. The simulation results of the proposed 5T SRAM cell
are discussed in Section 4. Last section is a conclusion and summary for the paper.
2. CONVENTIONAL 6T AND 5T SRAM CELL TOPOLOGIES
2.1 CONVENTIONAL 6T SRAM CELL
The conventional 6T SRAM cell is formed by a cross-coupled inverters (INV-1 and INV-2) and
two access transistors (MA1 and MA2), connecting the cell to the bit lines (BL and BLB), as
shown in Fig. 1 [12]. The cross-coupled inverters of the conventional 6T SRAM cell have two
stable states functioning to store either a logical ‘0’ or a logical ‘1’. The reading and writing
operations are achieved through the bit line BL and complementary bit line BLB of the memory
cell of Fig. 1. Transistors MP1 and MN1 are serially connected between power supply voltage
VDD and ground to form a first inverter INV-1 with a storage node A between the two transistors,
and, in a similar manner, transistors MP2 and MN2 are likewise connected between VDD and

3
ground to form a second inverter INV-2 with a storage node B. The gates of transistors of each
inverter are connected together and cross-coupled to the storage node of the other inverter. The
access transistors MA1 and MA2 are used to selectively couple or decouple the storage nodes (A
and B) from the corresponding bit lines. The gates of the access transistors MA1 and MA2 are
connected to respective word lines WL. The voltage on word line WL is delivered to the gates of
transistors MA1 and MA2 to control whether these transistors are switched on or off, thereby
coupling or decoupling the storage nodes from the bit lines.
Figure 1. Circuit diagram of conventional 6T SRAM cell.
It is important to note that, in write operation, complementary bit line BLB is pulled to zero using
write driver (not shown), while word line WL is asserted. Therefore, the access transistor MA2 is
turned on, which results in a voltage drop in the node B holding ‘1’. When this voltage falls
below VDD-VT(MP1), transistor MP1 starts the feedback action, wherein VT(MP1) is the threshold
voltage of transistor MP1. For stable write operation, transistor MA2 should be stronger than
transistor MP2. In read operation, read disturb may occur after the word line WL is asserted. The
voltage at the node A storing a ‘0’ slightly rises due to the voltage divider between the access
transistor MA1 and the driver MN1. If the voltage at node A rises above the threshold voltage of
transistor MN2, the cell may flip its state. In this case, stable read operation requires that MN1
should be stronger than MA1. Read stability failure increases with process variations, which
affect all the transistors in the cell.
It should be noted that since the conventional 6T SRAM circuit needs two bit lines which
individually consumes electric power, the power consumption required for the entire memory
array becomes large [13]. In general, in order to ensure the stable operation of the memory cell
circuit, the voltage level of the bit lines is pulled to a level near the power supply voltage VDD
before the start of read/write operations. This allows current to flow through the bit lines BL and
BLB, as such increases the power consumption. And thus, more power is consumed by the
memory cell circuit using two bit lines. Moreover, according to the memory cell circuit, current
always flows through one of the bit lines BL and BLB during the read operation whichever data is
held in the cross-coupled inverters. This also causes an increase in the power consumption [14].
2.2 CONVENTIONAL 5T SRAM CELL
Figure 2 is a circuit diagram of a conventional 5T SRAM cell [2]. As shown in Fig. 2, the access
transistor MA2 and bit line BLB in Fig. 1 have been removed to make up a five-transistor
configuration. The removal of such access transistor allows for an area savings up to 20-30%
compared to the conventional 6T SRAM cell, while its power consumption is substantially

4
reduced by one half [15]. Although the conventional 5T SRAM cells offer such significant
reduction in power consumption, a critical drawback is shown that in SRAM cells configured
with single-ended bit line, whenever a write operation is performed, a write failure may occur
[16-17]. In particular, it is relatively difficult to write a logical ‘1’ to a cell if the cell currently
stores a logical ‘0’. This is because when the bit line BL is logic high and the word line WL is
asserted, the transistors MA1 and MN1 fight one another. It is thus necessary to provide a method
of resolving write failures in five-transistor (5T) SRAM cells.
Figure 2. Circuit diagram of conventional 5T SRAM cell.
In order to resolve write ‘1’ issue of 5T SRAM cells, several techniques have been developed.
For example, boosting word line gate voltage [18-21], reducing the supply voltage VDD [4], [15],
[22-26], sizing cell transistors [15], [27-28], reduced bit line voltage [29-30], and raising the
source voltage VSS [31-34]. However, each of these techniques may cause a reduction in the drive
current of the transistors and in the operating speed of the cell, or has increased memory cell area
and a degradation in the manufacturing accuracy, or requires generation of a voltage above the
operating voltage, or requires a more complicated circuit design and more complicated device
process [14]. Hence, there is a need for an effective technique to improve the write-ability of 5T
SRAM cells which suffer from inability to write ‘1’.
Another problem with the 5T SRAM cells is that the data stored in the cells may be corrupted
when the cells are read [35]. This problem arises from the fact that a higher voltage on the bit line
is coupled to a lower voltage in the cell, causing the bit line voltage to drop and the cell voltage to
rise. To guarantee a correct write operation will occur, it is important to note that the storage node
A must be pulled up (or down) above (or below) the trip-voltage of INV-2 within the word line
WL is logic high, otherwise a write failure will occur. In more detail, writing a logical ‘0’ to a
cell, when initially storing a logical ‘1’, the high storage node A of the cell has to discharge the
bit line BL below the trip-voltage of INV-2. On the contrary, writing a logical ‘1’ to a cell, when
initially storing a logical ‘0’, the low storage node A of the cell must be pulled up by the pre-
charged bit line BL above the trip-voltage of INV-2. Undoubtedly, to write the wanted bit
properly in the cell, it may be necessary that the access transistor should be very conductive to
force the cross-coupled inverters to change its equilibrium condition [14]. Accordingly, the
SRAM cell should provide more reliable when the cell is written and less likely to be corrupted
when the cell is read. However, the access transistor should have a reduced conductivity for good
stability in reading and standby operations. These two requirements impose contradicting
requirements on cell transistor sizing.

5
3. THE PROPOSED 5T SRAM CELL
3.1 THE PROPOSED 5T SRAM CELL CONFIGURATION
The proposed 5T SRAM cell with voltage assist circuitries is illustrated in Fig. 3, which is formed
by a voltage control circuit, a pre-charging circuit, a standby start-up circuit and a word line
suppression circuit. Amongst them, the voltage control circuit is coupled to the sources
corresponding to the driver transistors of each row memory cells. This configuration is aimed to
control the source voltages of driver transistors under different operating modes. That is, the
circuit is employed to control the voltage of nodes L1 and L2 under different operating modes. In
the write mode, the voltage of node L1 (hereinafter, VL1) of the selected cell is set to VGS(M23),
wherein VGS(M23) is the gate-source voltage of transistor M23, and that of node L2 (hereinafter,
VL2) of the selected cell is set to the ground voltage. Thereby, in this manner, it can provide an
efficient solution to the writing ‘1’ issue and to improve write operations. In the read mode, a
two-stage read mechanism is introduced to speed up the reading speed and thus to avoid
unnecessary power consumption. In the first read stage, the voltage VL1 is set to a negative
voltage RGND to speed-up the reading speed. However, in the second read stage, the voltage VL1
is pulled up via transistor MN26 to the ground voltage to reduce power consumption. Under these
circumstances, the voltage RGND can effectively improve the reading speed without incurring
unnecessary power consumption even in technologies below 10nm. Finally, in the standby mode,
the voltage of nodes L1 and L2 are set to VGS(MN23) toreduce the leakage current.
MN24
MN25
RGND
Delay
D1
RC
S
L2
L1
S
MN22
MN21
MN23
MN26
S
VDD
P MP31
VDD
Delay
D2
MN41
MP41
S
INV
C
MP21
MP11
MN12
MN11
VDD
MP12
MN13
WL
BL
A
B
RC
MN52
WLC
MP51
MN51
WC
RC
GND GND
GND
WC
MN27
GND
word line
suppression
circuit
5T SRAM
cell
Standby
Start-up circuit
Pre-charging
circuit
Voltage control circuit
Figure 3. Circuit diagram of the proposed 5T SRAM cell.
Table 1 summaries the operating conditions under different operating modes. In Table 1, the write
control signal WC can be achieved by performing the AND operation on the write signal W and
its corresponding word line signal WL. Also, the read control signal RC can be achieved by
performing the AND operation on the read signal R and its corresponding word line signal WL. It

6
is worth noting that, in the non-read mode, the voltage of the read control signal RC is set to the
voltage RGND to prevent the leakage current caused by the transistor MN24.
Referring to Fig. 3, the pre-charging circuit is connected to the bit line BL in each column. The
function of the pre-charging circuit is to pull up the bit line BL of a selected column to power
supply voltage VDD before the start of read operation. And also, the standby start-up circuit is to
enable the SRAM cell to quickly switch to the standby mode, and thus effectively enhance the
standby performance.
Table 1: The operating conditions under different operating modes
RC WC S VL1 VL2 mode
RGND VDD 0 VGS(MN23) 0 write
VDD 0 0
RGND (1st stage)
0 (2nd stage)
0
read
RGND 0 VDD VGS(MN23) VGS(MN23) standby
RGND 0 0 0 0 hold
Referring again to Fig. 3, the word line suppression circuit is to provide a voltage of the selected
word line when the respective word lines are in an active state. Unlike the conventional 5T
SRAM cell in Fig. 2, a voltage VDD-VT(MN51) is applied to the word line control signal WLC of the
selected row cells so as to improve the cell read/write-ability, wherein VT(MN51) is the threshold
voltage of transistor MN51. In more detail, the word line suppression circuit weakens the access
transistor MN13 such that the voltage drop across the transistor MN13 increases and the voltage
drop between the transistor MN13 and the driver transistor MN11 reduces, thereby increasing the
read-ability. It is worth noting that the word line control signal WLC of the selected cell is
provided a voltage VDD-VT(MN51) during a read operation, however, the power supply voltage VDD
is provided during a write operation. In general, for a given cell size, a higher beta ratio improves
cell stability at the expense of lower access speed [17]. Lowering the word line voltage has the
effect of increasing the beta ratio.
In this suppression circuit, the read signal RC and the inverse write signal WC can be achieved
from the memory read/write control pin. When the signal RC is at logic high, it indicates that the
cell is in a read operation, and however the inverse write signal WC is at logic low indicates a
write operation. The voltage level of the signal WLCs under different operating modes is shown
in Table 2.
Table 2: Voltage level of the signal WLC under different operating modes
cells WL RC WC RC WLC mode
selected row cells VDD VDD VDD 0 VDD-VT(MN51) read
non-selected row cells 0 VDD VDD 0 0 read
selected row cells VDD 0 0 VDD VDD write
non-selected row cells 0 0 0 VDD 0 write
each cell 0 0 VDD VDD 0 non-access
It is worth noting that the paper introducing a two-stage reading mechanism to improve the
reading speed, as well as to avoid unnecessary power consumption. Furthermore, by using the
voltage word line suppression circuit to pull the voltage of the signal WLC in a selected row cells

7
down to VDD-VT(MN51) during a read operation, thereby to reduce the half-selected cells
disturbance. If the write control signal WC is at logic low, the voltage of node C (hereinafter, VC)
will be equal to that of the inverse standby control signal S . On the contrary, when the write
control signal WC is at logic high, voltage VC is the ground voltage. This allows stable write
operations. Furthermore, during the initial period in standby, the standby start-up circuit is
designed to rapidly charge the parasitic capacitance of node L1 to the voltage VT(MN23).
3.2 WRITE OPERATION
Refer to Fig. 3, before and during the write operation is performed, the standby start-up control
signal S is at logic low, thereby transistor MN26 is turned on, as such the voltage VL1 is pulled
down to ground. During the writing ‘0’ operation, the voltage of bit line BL (hereinafter, VBL) is
pulled down to logic low and the asserted word line WL turns on transistor MN13. Thus, node A
is at logic low and node B is at logic high. Conversely, during the writing ‘1’ operation, the
voltage VBL is pulled up to logic high and the asserted word line WL turns on transistor MN13.
Thus, node A is at logic high and node B is at logic low. In more detail, prior to the write
operation is performed, the write control signal WC is at logic low, transistor MP21 is turned on
and transistor MN27 is turned off. Thereby, the voltage VC is at logic high and thus to turn on
transistor MN26, as such the voltage VL1 is pulled down to the ground voltage. However, during
the write operation, the signal WC is at logic high, transistors MP21 is turned off and transistor
MN27 is turned on. Subsequently, the voltage VC is at logic low and thus to turn off transistor
MN26, as such the voltage VL1 is set to VGS(MN23). Thus, the issue concerning the difficulty of
writing ‘1’ can be resolved. Figure 4 shows the simplified circuit diagram during the write
operation.
Figure 4. Simplified circuit diagram during the write operation
.
The transients associated with a writing operation are detailed described below. Firstly, let us
consider the writing ‘0’ operation. Prior to the writing ‘0’ operation, the voltage VBL and that of
signal WLC are at logic low. During the writing ‘0’ operation, if a logical ‘0’ is stored previously,
the signal WLC transitions from a logic low to a logic high. As the signal WLC exceeds the
threshold voltage of transistor MN13 (hereinafter, VT(MN13)), transistor MN13 is turned on.
Subsequently, owing to the fact voltage VBL is at logic low, the voltage of storage node A
(hereinafter, VA) remains at the ground voltage. On the other hand, if a logical ‘1’ is stored
previously, the signal WLC transitions from a logic low to a logic high. As the signal WLC
exceeds the threshold voltage VT(MN13), transistor MN13 is turned on. Subsequently, owing to the

8
fact voltage VBL is at logic low, node A and node L1 will be discharged to ground until the end of
the writing ‘0’ operation.
Secondly, consider the writing ‘1’ operation. Prior to the writing ‘1’ operation, the signal WLC is
at logic low and the voltage VBL is at logic high. During the writing ‘1’ operation, if a logical ‘1’
is stored previously, the signal WLC transitions from a logic low to a logic high. When the signal
WLC exceeds the threshold voltage VT(MN13), transistor MN13 is turned on. Subsequently, owing
to the fact voltage VBL is at logic high and transistor MP11 remains on, the voltage VA will
remain at the power supply voltage VDD until the end of the writing ‘1’ operation. On the other
hand, if a logical ‘0’ is stored previously, the signal WLC transitions from a logic low to a logic
high. Subsequently, with the increase of signal WLC, the voltage VA will rise. As the signal WLC
exceeds the threshold voltage VT(MN13), transistor MN13 is turned on. Subsequently, owing to the
voltage VBL is at logic high and transistor MN11 remains on, and the voltage VB remains at a
voltage close to the power supply voltage VDD, transistor MP11 remains off. For a successful
write operation, it is desirable to pulling down the voltage VA (or VB) which has a stored value ‘1’
below the trip-voltage of the inverter. Meanwhile, the write initial transient voltage VAW of node
A must satisfy the following equation:
11 23
( 12)
11 13 23
MN MN
AW DD T MN
MN MN MN
R R
V V V
R R R

  
 
(1)
wherein VT(MN12) is the threshold voltage of the transistor MN12, RMN11, RMN13 and RMN23 are the
on-resistance of transistors MN11, MN13 and MN23, respectively. Consequently, the writing ‘1’
problem associated with the conventional 5T SRAM cell can be avoided. Now, transistor MN13
is still in the saturation region and transistor MN11 in the triode region. Although RMN13 may be
greater than RMN11, the NMOS diode MN23 can provide a voltage VGS(MN23) at node L1. As a
result, the voltage VA will rise up due to the voltage division along the driver and access
transistors. When the voltage exceeds a threshold, it causes the bit cell to flip due to regenerative
feedback. Hence, the writing ‘1’ operation is completed. Consequently, the writing ‘1’ problem
associated with the conventional 5T SRAM cell can be resolved. It is worth noting that the
voltage VL1 is VGS(MN23) when writing a logical ‘1’ to a logical ‘0’ is stored. After completing the
writing ‘1’ operation, the voltage VL1 will be discharged to ground via transistor MN26. It is
worth noting that the W/L ratio of transistor MN11 in Fig. 3 is designed smaller than that of
transistor MN1 shown in Fig. 1. Consequently, the writing ‘1’ problem associated with the
conventional 5T SRAM cell can be resolved.
3.3 READ OPERATION
As mentioned above, in the read operation, a two-stage reading mechanism is introduced to
increase the reading speed and thus to avoid unnecessary power consumption. Prior to a read
operation is performed, bit line BL is pre-charged to the power supply voltage VDD. Meanwhile,
the standby start-up control signal S, the write control signal WC and the read control signal RC
are at logic low, thereby transistors MP21 and MN25 are turned on and transistors MN24 and
MN27 are turned off, as such the voltage VC is at logic high and subsequently turn on the
transistor MN26. This leads to the voltage VL1 will be pulled down to ground. Figure 5 shows the
simplified circuit diagram during the read operation.
In the first reading stage, the read control signal RC is at logic high, thereby transistor MN24 is
turned on. At this time, since transistor MN25 would continue to conduct, the voltage VL1 will be
pulled down to a negative voltage RGND. Under this circumstance, the negative voltage RGND
can effectively improve the reading speed. Furthermore, in the second reading stage, the read
control signal RC remains at logic high and transistor MN24 remains on. Consequently, the

9
voltage VL1 is pulled up to the ground voltage due to transistor MN25 is turned off, and thus leads
to reduce unnecessary power consumption. It is note that the two-stage time interval is measured
as the time taken from a logic high on the read control signal RC to transistor MN25 is turned off.
This time interval can be adjusted by the falling time of the inverter INV and the delay time of the
delay circuit D1. In addition, during the two-stage read operation transistor MN26 is always on.
The transients associated with a reading operation are detailed described below. Firstly, before the
reading ‘1’ operation, transistor MN11 is off and transistor MN12 is on, the voltages VA and VB
are at VDD voltage level and the ground voltage, respectively. The voltage VBL is equal to the
power supply voltage VDD due to the pre-charging circuit. During the read operation, since the
voltage VWLC is at VDD-VTMN51 voltage level, transistor MN13 is on. Thereby the voltage VBL can
be effectively kept at VDD until the end of the reading ‘1’ operation. It is worth noting that, in the
first stage, the voltage VL1 can be expressed as:
26
1
24 25 26
MN
L
MN MN MN
R
V RGND
R R R
 
 
(2)
wherein RGND is the acceleration read voltage (RGND) which is a negative voltage, RMN24,
RMN25 and RMN26 are the on-resistance of transistors MN24, MN25 and MN26, respectively.
While, in the second stage, the voltage VL1 can be expressed as:
1
L
V GND
 (3)
As a result, the unnecessary power consumption can be reduced effectively. Furthermore, in order
to effectively reduce the half-selected cell disturbance and the leakage current during the reading
‘1’ operation, the absolute value of the voltage RGND may be set to be lower than the voltage
VT(MN11) in reading ‘1’, i.e.,
( 11)
| | T MN
RGND V
 (4)
Wherein, |RGND| denotes the absolute value of the voltage RGND and VT(MN11) is the threshold
voltage of transistor MN11.
Figure 5. Simplified circuit diagram during the read operation.

10
Secondly, consider the reading ‘0’ operation. Before the read ‘0’ operation is performed,
transistor MN11 is on and transistor MN12 is off, voltage VA and voltage VB are ground and VDD,
respectively. And also, the voltage VBL is equal to VDD due to the pre-charging circuit. During the
reading ‘0’ operation, since the voltage VWLC is at VDD-VT(MN51) voltage level, transistor MN13 is
turned on. Meanwhile, the initial transient voltage VAR of node A must satisfy the following
equation:
11 24 25 26
13 11 24 25 26
11 13 26 13
24 25 11 13 26 11 13
( 12)
( ) ||
( ) ||
( ) ||
+
( )||
MN MN MN MN
AR DD
MN MN MN MN MN
MN MN MN MN
MN MN MN MN MN MN MN
T MN
R R R R
V V
R R R R R
R R R R
RGND
R R R R R R R
V
 
 
  

 
   

(5)
Where in, VAR is the initial transient voltage of node A, VTMN12 is the threshold voltage of
transistor MN12, RMN11, RMN13, RMN24, RMN25 and RMN26 are the on-resistance of transistors MN11,
MN13, MN24, MN25 and MN26, respectively. It is worth noting that, the voltage RGND is
designed to be lower than the ground voltage and its absolute value is designed to be lower than
the voltage VT(MN11). Furthermore, during the read ‘0’ operation, the voltage VWLC is set to a VDD-
VT(MN51) voltage level, as such the on-resistance of transistor MN13 can be increased to satisfy the
equation (5) and can reduce the half-selected cell disturbance in reading ‘0’ operation.
3.4. STANDBY OPERATION
Refer again to Fig. 3, prior to the standby operation is performed, the standby control signal S is
at logic low, and thus, transistor MP41 is turned off and transistor MN41 is turned on. While
during the standby operation, since the control signal S is at logic high, transistor MP41 is turned
on and transistor MN21 is turned off. In addition, the logic high signal S is also to turn on
transistor MN22 which acts as an equalizer. Consequently, with the conduction of transistor
MN22, both the VL1 and VL2 are equal to the threshold voltage of transistor MN23 (VT(MN23)). It is
worth mentioning that node L1 can be rapidly charged to VT(MN23) at the initial period of the
standby mode due to transistor MN41 remains on, and thereby improving the standby efficiency.
Note that the initial period is determined as the time taken from a logic high on the signal S to the
state of transistor MN41 is turned off. This time interval can be adjusted by the delay time of
delay circuit D2. It is worth noting that after the initial period of the standby mode, transistor
MN41 is turned off and no current flows. Figure 6 shows the simplified schematic of the
proposed design during the standby mode.
Figure 6. Simplified circuit diagram during the standby operation.

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4. SIMULATION RESULTS
To evaluate performance, different SRAM cell structures discussed in this paper were simulated
using a 90nm CMOS technology. All simulations were carried out at nominal conditions:
VDD=1.2V and at room temperature. In conventional 5T SRAM cell shown in Fig. 2, access
transistor MA1 is less conductive than driver transistor MN1, thereby making it more difficult to
write a logical ‘1’ to cell over a logical ‘0’ stored. Figure 7(a) shows the simulated waveform of a
writing ‘1’ failure. It is worth noting that sizing of MN11 should ensure that inverter INV-2 does
not switch causing a destructive read.
To elucidate the improvement in writing ‘1’ issue, suppose that node A stores ‘0’ and it needs to
change to ‘1’ during a write cycle. Before the write operation is performed, both transistors MP11
and MN12 are off and both transistors MP12 and MN11 are on. When the writing ‘1’ is
performed, if the voltage of word line WL (VWL) exceeding the threshold voltage of transistor
MN13 (VT(MN13)), transistor MN13 will be turned on. Upon write operation, voltage VBL remains
at the pre-charge level as such transistor MN11 remains on and transistor MP11 off. Effectively,
transistors MN11 and MN13 make up a voltage divider and have been connected to the input of
INV-2, whose output is now no longer at zero. Meanwhile, node A can be charged toward
11 11 13
( / ( ))
DD MN MN MN
V R R R
  , where RMN11 is the on-resistance of transistor MN11, and RMN13 is
the on-resistance of transistor MN13, respectively. Traditionally, since the W/L ratio of transistor
MN11 is designed smaller than that of transistor MN1 shown in Fig. 1, there will be no
destructive reading occurs. In addition, it enables the voltage of node A (VA) higher than the
threshold voltage of transistor MN12 (VT(MN12)). Consequently, transistor MN12 is on and node B
is discharged to a lower voltage level. And then, the voltage VA rises since transistor MN11 is
now possess higher resistance value. This higher resistance value helps pulling up node A toward
much higher voltage level. Hereafter, by using of INV-2, this much higher voltage VA will pull
node B down to much lower voltage level. Furthermore, by using of INV-1, this much lower
voltage level on node B will pull VA up to much higher voltage. In this way, such operations are
alternately repeated until VA reaches VDD. Hence, the writing ‘1’ operation of the proposed cell is
accomplished. The simulated waveform of a successful writing in the proposed 5T SRAM cell is
shown in Fig. 7(b). It is evident that the proposed 5T SRAM cell provides an efficient solution to
the writing ‘1’ issue, that is, the proposed 5T SRAM cell enabling a logical ‘1’ to be easily
written to the SRAM cell, as compared to the conventional 5T SRAM cells.
(a) (b)
Figure 7. (a) Transient waveforms of a write failure in the conventional 5T SRAM cell, (b) Transient
waveforms of a successful writing in the proposed 5T SRAM cell.
Upon read operation, due to a two-stage reading mechanism is introduced to improve the reading
speed as well as to avoid unnecessary power consumption. During the first stage of the read
operation, the voltage VL1 is set to the acceleration read voltage RGND which is lower than the
ground voltage, as shown in Fig. 8. As such to effectively increase the reading speed, and the
voltage VL2 is set to the ground voltage. In the second stage, the voltages VL1 and VL2 are set to
0→0 0→0
0→1 0→1 1→1
1→1 1→0
1→0

12
the ground voltage to reduce unnecessary power consumption. Table 3 shows the reading speed
for the proposed design and the conventional 6T SRAM cell in different corner models. An
advantage of the proposed design over the conventional 5T SRAM cell is that it is unnecessary to
boost the word line signal above VDD to speed up the read operation. Furthermore, this design has
the additional advantage of increased current through the driver transistor during a read operation,
and consequently lower read delay.
Figure 8. Signal waveforms during a read operation.
Table 3. Reading speed comparisons
Corner
model
Conventional 6T
SRAM (ns)
Proposed 5T SRAM
(ns)
Improvement
(%)
TT 0.1205 0.0748 37.9
SS 0.1548 0.1052 32.0
FF 0.1985 0.132 33.5
It can be seen from Table 3, compared with the conventional 6T SRAM cell in different corner
models, the reading speed of the proposed design is significantly speed-up to 37.9%, 32.0% and
33.5%, respectively.
Finally, upon standby mode shown in Fig. 6, the voltage VA remains at VT(MN23), the voltage VWL
is set to the ground voltage and the voltage VBL is set to VDD, respectively. Therefore, the gate-
source voltage VGS of transistor MN13 is negative. In contrast, the voltage VGS of transistor MA1
in Fig. 1 is equal to zero. For NMOS transistors, according to the GIDL effect, the sub-threshold
current at VGS=-0.1 is approximately 1% of that at VGS=0. Accordingly, the leakage current I1
flows through transistor MN13 caused by the GIDL effect is much smaller than that of flowing
through transistor MA1 in Fig. 1. Furthermore, the drain-source voltage VDS of transistor MN13
is VDD-VTMN23, whereas the voltage VDS of transistor MA1 in Fig. 1 is VDD. According to the
DIBL effect, the leakage current I1 flowing through transistor MN13 is also less than that of
flowing through transistor MA1 in Fig. 1. As a result, the leakage current flows through transistor
MN13 is much smaller than that flowing through transistor MA1 in Fig. 1. Next, the source-drain
voltage VSD of transistor MP11 is VDD-VTMN23 in contrast to the VSD = VDD of transistor MP1 in
Fig. 1. According to the DIBL effect, the leakage current I2 flowing through transistor MP11 will
be less than that of flowing through transistor MP1 in Fig 1. Thus, the base-source voltage VBS of
transistor MN12 is negative, and the drain-source voltage VDS of transistor MN12 is VDD-VTMN23.
On the contrary, the VBS of transistor MN2 in Fig. 1 is zero, and the VDS of transistor MN2 is
VDD. According to the body effect and DIBL effect, the leakage current I3 flows through

13
transistor MN12 is much smaller than that of flowing through transistor MN2. From the above
analysis, it can be seen that the proposed 5T single-port SRAM having a lower leakage current
compared with the conventional 6T SRAM. Table 4 shows the standby leakage current for the
proposed design and the conventional 6T SRAM cell in different corner models.
Table 4. Leakage current comparisons
Corner
model
Proposed 5T SRAM
(pA)
Conventional 6T SRAM
(pA)
Improvement
(%)
TT 2.0536 22.2203 90.8
SS 1.0587 1.6191 34.6
FF 38.3415 309.2402 87.6
As it can be seen from Table 4, compared with the conventional 6T SRAM cell in different corner
models, the standby leakage current of the proposed design is significantly reduced 90.8%, 34.6%
and 87.6%, respectively.
5. CONCLUSIONS
In this paper, a new 5T single-port SRAM cell with voltage assist is proposed. Firstly, a word line
suppression circuit is designed to provide a gate voltage of access transistor of the selected word
line lower than the power supply voltage by a threshold voltage, as such to improve the cell
read/write-ability. Furthermore, in order to resolve the writing ‘1’ issue, the voltage VL1 of the
selected cell is set to a gate-source voltage VGS(M23) and the voltage VL2 is set to the ground
voltage, and consequently fast writing also can be achieved. In addition, in the read operation, a
two-stage read mechanism is introduced to speed up the reading speed and to avoid unnecessary
power consumption. To speed-up the reading operation, a voltage control circuit facilitates such a
read operation by supplying the SRAM cell with a speed-up reading voltage RGND that is lower
than the ground voltage during a first-stage read operation. However, during a second-stage read
operation, the voltage VL1 is pulled up to the ground voltage to reduce unnecessary power
consumption. Finally, in the standby mode, the voltages VL1 and VL2 are set to a gate-source
voltage VGS(MN23) toreduce the leakage current.
An advantage of the proposed design over the conventional SRAM cell is that it is unnecessary to
boost the word line signal above the power supply voltage VDD to speed up the read operation.
This design has the additional advantage of increased current through the driver transistor during
a read operation, and thus decreases read delay. Simulation results for the proposed 5T cell design
confirm that there is conspicuous improvement over the conventional 6T SRAM cell while it
allows writing ‘1’ on the cell with write assist. In addition, with the proposed voltage assist leads
to a 37.9%, 32.0% and 33.5% reading speed-up in different corner compared with the
conventional 6T SRAM cell. Finally, with the proposed design leads to a 90.8%, 34.6% and
87.6% less standby leakage in different corner compared with the conventional 6T SRAM cell.
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AUTHORS
Chien-Cheng Yu received the B.S. and M.S. degrees from Feng Chia University and National Taiwan
Normal University, respectively. He is currently working toward the Ph.D. degree in electrical engineering
at National Chung Hsing University, Taichung City, Taiwan. In 1986, he joined Hsiuping University of
Science and Technology, Taichung City, Taiwan, where he is now an Associate Professor. His current
research interests include design and analysis of high-speed, low-power integrated circuits and low-voltage
low-power embedded SRAM circuit design.
Ming-Chuen Shiau received the B.S., M.S. and Ph.D. degrees in Electronic Engineering from National
Chiao Tung University, Hsinchu, Taiwan. He is currently a Professor with the Department of Electrical
Engineering at Hsiuping University of Science and Technology, Taichung City, Taiwan. His current
research interests include low-power digital circuit design, SRAM design and low-voltage embedded
memory circuit design.
Ching-Chih Tsai received the Diplomat in electrical engineering from National Taipei Institute of
Technology, Taipei, Taiwan, in 1981, the M.S. degree in control engineering from National Chiao Tung
University, Hsinchu, Taiwan, in 1986, and the Ph.D. degree in electrical engineering from Northwestern
University, Evanston, IL, in 1991. He is currently a Distinguished Professor in the Department of Electrical
Engineering, National Chung Hsing University, Taichung City, Taiwan. His current research interests
include ultrasonics, mechatronics, mobile robotics, advanced control methods, embedded control systems,
and their applications to mobile robots and industrial control systems.

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