High Speed and Low Pedestal Error Bootstrapped CMOS Sample and Hold Circuit

International Journal of Electrical and Computer Engineering (IJECE)
Vol. 8, No. 6, December 2018, pp. 4148∼ 4156
ISSN: 2088-8708, DOI: 10.11591/ijece.v8i6.pp4148-4156 4148
High Speed and Low Pedestal Error Bootstrapped CMOS
Sample and Hold Circuit
Agung Setiabudi1
, Hiroki Tamura2
, and Koichi Tanno3
1
Department of Materials and Informatics, University of Miyazaki, Japan
2
Department of Environmental Robotics, University of Miyazaki, Japan
3
Department of Electrical and System Engineering, University of Miyazaki, Japan
Article Info
Article history:
Received Jan 23, 2018
Revised May 31, 2018
Accepted Jun 20, 2018
Keywords:
Sample and Hold Circuit
Error Reduction
CMOS Switch
Clock Feedthrough
Channel Charge Injection
Bootstrap Circuit
Abstract
A new high speed, low pedestal error bootstrapped CMOS sample and hold (S/H) circuit
is proposed for high speed analog-to-digital converter (ADC). The proposed circuit is
made up of CMOS transmission gate (TG) switch and two new bootstrap circuits for
each transistor in TG switch. Both TG switch and bootstrap circuits are used to decrease
channel charge injection and on-resistance input signal dependency. In result, distortion
can be reduced. The decrease of channel charge injection input signal dependency also
makes the minimizing of pedestal error by adjusting the width of NMOS and PMOS
of TG switch possible. The performance of the proposed circuit was evaluated using
HSPICE 0.18-µm CMOS process. For 50 MHz sinusoidal 1 V peak-to-peak differen-
tial input signal with a 1 GHz sampling clock, the proposed circuit achieves 2.75 mV
maximum pedestal error, 0.542 mW power consumption, 90.87 dB SNR, 73.50 SINAD
which is equal to 11.92 bits ENOB, -73.58 dB THD, and 73.95 dB SFDR.
Copyright © 2018 Institute of Advanced Engineering and Science.
All rights reserved.
Corresponding Author:
Koichi Tanno,
Department of Electrical and System Engineering, University of Miyazaki,
1-1 Gakuenkibanadai-nishi, Miyazaki, 889-2192, Japan.
tanno@cc.miyazaki-u.ac.jp
1. INTRODUCTION
Analog techniques have dominated signal processing for years, but digital techniques are slowly en-
croaching into this domain. Digital signal processing (DSP) is becoming popular because its flexibility to per-
form various processing operations. This technology is widely used in many different domains, such as wireless
communications, medical electronics, measurement instrumentation, digital multimedia, etc. One of the essen-
tial component of DSP is analog to digital converter (ADC), because most natural signals in the world (such as
voltage, current, temperature and pressure) are analog. ADC perform the digitalization of analog signals at fixed
time period, which is generally specified by the application. Since the digital signal that will be processed by
DSP is originally from ADC, performance of the DSP is highly dependent to the performance of ADC itself. To
avoid premature signal degradation, the ADC must achieve specified sampling speed, resolution and precision.
To meet the specifications, several techniques and ADC architectures have been proposed. Parallel (flash) ADC
is by far the fastest and conceptually simplest has been reported [1]. The drawbacks of this architecture are
the resolution is limited by circuit complexity, high power dissipation, and comparator and reference mismatch.
To reduce hardware complexity, power dissipation and die area, and to increase the resolution but to maintain
high conversion rates, flash converters can be extended to pipeline [2], delta-sigma ADC [3, 4], or successive
approximation ADC [5].
The key building block in front-end of ADC is Sample and Hold circuit (S/H). This front-end circuit
is almost inevitable in some types of ADC. The main function of S/H is to take analog input signal samples
and hold its value until ADC can process the information. In other word, the accuracy and the speed of the
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IJECE ISSN: 2088-8708 4149
converter is strongly dependent on performance of S/H, because this circuit provide the information which will
be processed by the converter. The architectures of S/H can roughly be divided into two, open-loop and closed-
loop architectures. The main difference between them is that in closed-loop architectures, the hold capacitor
is placed in a feedback loop, at least in hold mode, whereas the open-loop one has no feedback. Moreover,
the characteristics between these two architectures are mostly different. The open-loop architecture offer the
simplicity and speed solution, however its accuracy is limited by distortion arising from nonlinearity of the
switch which is caused by its signal dependent on-resistance and pedestal error which is caused by the signal
dependent channel charge injection and clock feedthrough. On the other hand, enclosing the capacitor in the
feedback loop on the close-loop architecture can reduces the effect of nonlinearity and signal-dependent charge
injection and clock feedthrough from the MOS switches, offering better accuracy as a result. Unfortunately, an
inevitable consequence of the use of feedback is conditionally stability problem. Furthermore, since close-loop
architecture uses Op-Amp, its bandwidth and slew rate directly limit the speed of the S/H. The complexity and
power consumption of the Op-Amp also rise another problem.
In order to maintain the simplicity and the speed, and to increase the accuracy of S/H, in this paper a new
open-loop CMOS S/H is proposed. The proposed circuit consists of new bootstrap circuit to overcome channel
charge injection and clock feedthrough signal dependent problem, and transmission gate (TG) switch to deal with
on-resistance problem. In this paper, a simple theoretical and numerical method to minimize the pedestal error is
also presented. The rest of this paper is organized as follows. Section 2 explain the operation of S/H, especially
open-loop S/H. In this section, channel charge injection and clock feedthrough mechanism will be reanalyzed
in detail. Several previous existing techniques related to open-loop S/H will also be presented in this section.
Section 3 presents the proposed circuit and its theoretical analysis. Section 4 presents the simulation result of
proposed circuit and the comparison with other works. At last, the conclusion is presented in section 5.
2. OPEN-LOOP SAMPLE AND HOLD CIRCUIT
Simple open-loop S/H circuit consists of two main part, ﬁrst part is analog switch and second one is
hold capacitor. The analog switch is used to control the connection between signal-source (input) node and data-
holding (output) node, whereas the hold capacitor is used to hold the data in output node. The operation of S/H
circuit takes place in two phases: sampling and hold. When the clock voltage (φ) which is applied to the gate of
transistor is high, transistor is on. Channel is appearing underneath the gate and connecting drain and source of
the transistor. In this phase, input node and output node is connected and sampling function is performed. After
the switch is turned off, the data appearing in the holding (output) node will be held until the next operation step
occurs, this phase is called hold phase. In the transition phase between sampling and hold, the channel charge
disappears through either the source/drain electrodes or substrate electrodes. Charge which disappears through
source/drain is deposited on hold capacitor creating an error component to the sample voltage. This phenomenon
is called channel charge injection. In an MOS transistor it is also known that it has parasitic capacitance that
is formed by overlapping between gate and diffusion (source and drain). When transistor turns off, this overlap
capacitance also ﬂows the charge to hold capacitor creating another error component to sample voltage. This
phenomenon is called clock feed through. These two mechanisms are the main sources that create an error in
sample voltage. Some researches have been done and published regarding these two mechanisms[6, 7].
2.1. Channel Charge Injection, Clock Feedthrough and On-Resistance Problem
In the sampling phase, transistor is on and a channel exists at the oxide-silicon interface. This phase
can be depicted in Figure. 1. (a). Assuming Vin = Vout, the total charge in the inversion layer (channel) can be
obtained as
Qch = −WLCox(φh − Vin − Vt) (1)
Qch = −WLCox(Vdd − Vin − Vt) (2)
where W is channel width, L is channel length, Cox is oxide capacitance per unit area, φh is the high level of
clock voltage, and Vt is threshold voltage of the transistor. In many application the high level of clock voltage is
equal to supply voltage (φh = Vdd). Thus, the equation (1) can be rewritten as (2).
High Speed and Low Pedestal Error Bootstrapped CMOS... (Agung Setiabudi)

4150 ISSN: 2088-8708
(a)
(b)
Figure 1. S/H phases: (a) Sampling phase of S/H circuit, (b) On-off transition phase of S/H circuit
When the switch turns off (S/H circuit enter hold phase), Qch exits through the source and drain terminals
like shown in Figure. 1. (b). The charge injected to left side of Figure. 1. (b) is absorbed by the input source,
creating no error. Whereas, the charge injected to the right side is deposited on CH, creating an error in the output
node. Assuming that the amount of charge which flow to the left side equal to the one which flow to the right
side, it is obtained that the amount of charge causes error in the output node is Qch/2. Therefore, the error voltage
caused by channel charge injection then can be written as follow
∆Vcci = −
WLCox(Vdd − Vin − Vt)
2CH
(3)
On the turning off process, the MOS switch also couples the clock transition to the hold capacitor (CH)
through this overlap capacitance (gate-drain or gate-source overlap capacitance). The coupling current that flow
from gate to drain/source through overlap capacitance causes an error in the output node. The error voltage
caused by clock feedthrough can be written as follow
∆Vcft = −
LovWCox
LovWCox + CH
φh (4)
Where Lov is gate-drain or gate-source overlap length. These errors are the main source of accuracy
problem in open-loop S/H. Error caused by channel charge injection is more dominant compared with error
caused by clock feedthrough, because the length of overlap capacitance in eq. (4) is very small. In some cases,
this error is often neglected. The other problem that limit the accuracy of open-loop S/H is nonlinearity. This
nonlinearity is caused by signal dependent on-resistance shown in eq. (5). Where µ is mobility of the electron.
From the equation, it also can be inferred that the switch has narrow input swing. because as the input signal
equal to Vdd − Vt, the on-resistance become infinity.
Ron =
1
µCox
W
L
(Vdd − Vin − Vt)
(5)
IJECE Vol. 8, No. 6, December 2018 : 4148 – 4156

IJECE ISSN: 2088-8708 4151
2.2. Open-Loop S/H Circuit Using TG Switch
The simple way to decrease the error and nonlinearity caused by channel charge injection, clock feedthrough
and non-uniform on-resistance of the switch is by using TG switch. Figure 2 shows schematic of basic S/H cir-
cuit using CMOS switch. The switch part of this circuit consist of two MOS transistors, N type MOS transistor
and P type MOS transistor (TG switch). By using TG switch, the problem of narrow input voltage swing and
non-uniform on-resistance can be solved. Input range of this switch is nearly 0 V - Vdd, on-resistance of this
switch is also relatively uniform compared with single transistor switch. On-resistance of TG switch is written in
eq. (6).
Ron =
1
µnCox
Wn
Ln
(Vdd − Vin − Vtn) + µpCox
Wp
Lp
(Vin − |Vtp|)
(6)
Figure 2. S/H circuit using TG switch
Since the TG switch is built using two type MOS transistors, the error in this switch is also caused by
those two transistor. And as well-known that the characteristic of N type and P type MOS transistor are mutually
opposite. Thus, the error of NMOS and PMOS are mutually compensate. The total error in this S/H circuit is
written in eq. (7).
∆V =
1
2CH
Cox[WpLp(Vin − |Vtp|) − WnLn(Vdd − Vin − Vtn)]
+
LovpWp − LovnWn
LovnWnCox + LovpWpCox + CH
Coxφh
(7)
Where subscript n and p refer to N type and P type MOS transistor, respectively. The equations used for
modeling hold error in this paper is basic model, neglecting body effect and gate/clock voltage falling rate. The
ﬁrst term in this equation is the error caused by channel charge injection. This term is dominant term. The second
term is error caused by clock feedthrough. This term is less dominant and can be neglected in some application
with less accuracy consideration.
Theoretically, the performance of open-loop S/H circuit using TG switch is better than the other one
with single MOS transistor. However, in the high speed and high accuracy application, the use of open-loop S/H
circuit with TG switch is not recommended. Because the effect of channel charge injection, clock feedthrough
and on-resistance signal dependent is still very signiﬁcant to accuracy degradation. To overcome this problem,
many researchers develop and propose new techniques. The use of dummy transistor to eliminate the effect
of channel charge injection and clock feedthrough has been reported. However, the performance of dummy
transistor is limited by the transistor mismatch and unbalance impedance at drain and source terminal of the
switch. The other technique is bottom plate sampling. Recently, the improvement of bottom plate sampling is
reported in [8, 9]. This technique shows decent performance to decrease charge injection and clock feedthrough
effect, but the complex clocking phase in this technique rises another problem in system complexity. Several
other techniques also have been reported, such as the use of current conveyor as analog switch, current controlled

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current mirror switching, distributed S/H, substrate-biasing-effect attenuated T switch [10], cross couple switch
capacitor network, offset cancellation replica circuit [11], bootstrapping techniques [12], and other techniques
[13, 14, 15, 16]. For open-loop S/H circuit the bootstrapping technique is a suitable one, this technique keeps the
gate-source voltage of sampling transistor ﬁx at particular value. This approach keeps the on-resistance constant
and thus improve the switch linearity. By keeping the gate-source voltage ﬁx, it also decreases the channel charge
injection and clock feedthrough signal dependent.
3. PROPOSED S/H CIRCUIT
To solve the problems presented in section 2., a new bootstrapped S/H circuit with TG switch is pro-
posed. As depicted in Figure. 3, unlike the common bootstrap circuit [12], the proposed circuit is made up of
CMOS TG switch and novel bootstrap circuit structure for each transistor in TG switch (NMOS bootstrap circuit
and PMOS bootstrap circuit). The proposed circuit is operated in differential mode, but for simplicity, Figure. 3
shows only single input mode circuit. CMOS TG switch is formed by Ms1 and Ms2, while C1, M1−5 and C2,
M6−10 form NMOS and PMOS bootstrap circuits, respectively.
Figure 3. Proposed S/H circuit
These circuits are controlled by φ and ¯φ which are non-overlapping clocks with opposite phases. The
operation principles of the proposed circuit is explained as follows. When φ is high, (φ = 1, ¯φ = 0), in NMOS
bootstrap circuit, M1−3 are on and M4−5 are off, cause Vdd is applied at top plate of capacitor C1, and since M2
is on, the output of this circuit which is connected to the gate of NMOS in TG switch is connected to ground
(Vgn = 0). While in PMOS bootstrap circuit, M6−8 are on and M9−10 are off. Thus, Vdd is applied at top plate
of capacitor C2, and since M7 is on, the output of this circuit which is connected to the gate of PMOS in TG
switch is connected to Vdd (Vgp = Vdd). In this phase, the circuit is in hold phase as depicted in Figure. 4.
(a) (b)
Figure 4. Hold phase in bootstrap circuit: (a) NMOS bootstrap, (b) PMOS bootstrap

IJECE ISSN: 2088-8708 4153
When φ is low, (φ = 0, ¯φ = 1), in NMOS bootstrap circuit, M1−3 are off and M4−5 are on, cause the
bottom plate of capacitor C1 connects to Vin and its top plate connects to Vgn. Thus, the output of this circuit,
Vgn equal to Vin + Vdd. While in PMOS bootstrap circuit, M6−8 are off and M9−10 are on, cause the top plate
of capacitor C2 connects to Vin and its bottom plate connects to Vgp. Therefore, the output of the circuit, Vgp
equal to Vin −Vdd. In this phase, the circuit is in sampling phase as depicted in Figure. 5. Based on the operation
principles explanation, the output of NMOS and PMOS bootstrap circuits can be written in Eqs. (8) and (9),
respectively.
Vgn =
Vin + Vdd, in sampling phase
0, in hold phase
(8)
(a) (b)
Figure 5. Sampling phase in bootstrap circuit: (a) NMOS bootstrap, (b) PMOS bootstrap
Vgp =
Vin − Vdd, in sampling phase
Vdd, in hold phase
(9)
By using Eqs. (8) and (9), the on-resistance and the total error of the proposed circuit can be written in
Eqs. (10) and (11), respectively.
Ron =
1
µnCox
Wn
Ln
(Vdd − Vtn) + µpCox
Wp
Lp
(Vdd − |Vtp|)
(10)
∆V =
1
2CH
Cox[WpLp(Vdd − |Vtp|) − WnLn(Vdd − Vtn)]
+
LovpWp (2Vdd − Vin) − LovnWn (Vdd + Vin)
LovnWnCox + LovpWpCox + CH
Cox
(11)
Equation (10) shows that the on-resistance of proposed S/H circuit is independent to input signal. Al-
though it is well known that the threshold voltage of transistor is input signal dependent, it can be neglected
because its value variation is small enough. By this condition, the linearity of S/H can be increased and the
distortion can be reduced. This equation also shows that the overdrive voltage of each transistor in TG switch
increase by Vin. It means the value of on-resistance becomes smaller and it can increase the switching speed.
Equation (11) shows that the total error dominant term (error caused by channel charge injection) is
independent to input signal Vin. Although the second term (error caused by clock feedthrough) becomes in-
put signal dependent, this technique still gives the advantages. Because the input signal dependent shifts from
dominant term to less dominant term, so that the total input signal dependent can decrease.
4. SIMULATION RESULTS
The performance of the proposed circuit was evaluated using HSPICE with 1P, 5M, 3-well, 0.18-m
CMOS process (BSIM3v3.2 LEVEL53). Figure 6 shows the input and output waveform of bootstrap circuit for a
sinusoidal input of 1 V peak-to-peak at 50 MHz with a 1 GHz sampling clock. From this simulation result it can

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be known that in the sampling phase the circuit can track the input (dashed line) voltage and give the output (solid
line) of Vin + Vdd for NMOS bootstrap circuit and Vin − Vdd for PMOS bootstrap circuit. Figure 7 (a) shows
the transient response of the proposed S/H circuit for 50 MHz sinusoidal 1 V peak-to-peak differential input
signal with a 1 GHz sampling clock. The common mode (CM) level of differential signal is 0.25 V. The pedestal
error as the function of differential input voltage is shown in Figure. 7 (b). For 50 MHz sinusoidal 1 V peak-to-
peak differential input signal with a 1 GHz sampling clock, the maximum absolute value and root-means-square
(RMS) value of pedestal error are 2.75 mV and 1.72 mV, respectively with 0.542 mW power consumption.
(a) (b)
Figure 6. The input and output waveform of bootstrap circuit for a sinusoidal input of 1 V peak-to-peak at 50
MHz with a 1 GHz sampling clock: (a) NMOS bootstrap circuit, (b) PMOS bootstrap circuit
(a) (b)
Figure 7. Simulation results of proposed S/H circuit for 50 MHz sinusoidal 1 V peak-to-peak differential input
signal with a 1 GHz sampling clock: (a) Input and output waveform, (b) Pedestal error
Figure 8 shows the sampled signal spectrum of 50 MHz sinusoidal 1 V peak-to-peak differential input
signal with a 1 GHz sampling clock. From the analysis, in the Nyquist bandwidth the proposed circuit has
performances of 90.87 dB SNR, 73.50 SINAD/SNDR which is equal to 11.92 bits ENOB, -73.58 dB THD, and
73.95 dB SFDR. Furthermore, the simulation of the circuit in various sampling frequencies and input frequencies
are done. The input frequency varies from 10 MHz to 50 MHz in the step of 10 MHz while the sample frequency
varies from 300 MHz to 1 GHz in the step of 100 MHz. From this condition the proposed circuit has average
performances of 69.64 dB SNR, 64.50 SINAD/SNDR which is equal to 10.42 bits ENOB, -77.59 THD, and 69.64
dB SFDR. At last, a comparison of main performance of the proposed circuit with other works is summarized in
Table. 1. In general, the proposed circuit shows better performance than other work. Compared with the same
bootstrapping technique [12], the proposed circuit also shows better SNDR and higher sampling rate.

IJECE ISSN: 2088-8708 4155
Figure 8. Sampled signal spectrum of 50 MHz sinusoidal 1 V peak-to-peak differential input signal with a 1 GHz
sampling clock
Table 1. Comparison of main performance of the proposed circuit with other works
Parameter Proposed [11] [13] [14] [15] [16] [10] [9] [8] [12]
Sampling rate (MS/s) 1000 500 40 200 330 200 100 500 250 280
Input frequency (MHz) 50 10 - 40 80 - 10 220 20 7
Input Amplitude (V) 1 0.8 1.4 0.8 1.2 2 2 1.6 1.6 0.5
SNR (dB) 90.87 - 67 - - - - - - -
SINAD/SNDR (dB) 73.50 60.50 - 45 - - 85.5 76 - 57.01
ENOB (bits) 11.92 9.8 9 7.2 11 12 13.9 12.33 14 9.18
THD (dB) -73.58 -60.5 -56 - -68.3 - - - - -
SFDR (dB) 73.95 69 57 60 - 87 92.87 - 80 -
Pedestal Error (mV) <2.75 <4.9 - - <0.8 - - - - -
Power consumption (mW) 0.542 6 0.5 22 26.4 - - - - 0.88×10−3
CMOS Technology (nm) 180 90 130 180 350 350 130 350 350 90
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