INVERTING BUCK-BOOST DCDC CONVERTER DESIGN CHALLENGES

International Journal of Embedded Systems and Applications (IJESA) Vol.4, No.1, March 2014
DOI : 10.5121/ijesa.2014.4101 1
INVERTING BUCK-BOOST DCDC CONVERTER DESIGN
CHALLENGES
Karim El khadiri1
and Hassan Qjidaa2
1,2
SidiMouhamed Ben Abdellah University,DharMahraz Science Faculty, Fez,Morocco
ABSTRACT
This paper presents an inverting buck-boost DCDC converter design. A negative supply voltage is needed
in a variety of applications, but only a few DCDC converters are available on the market. OLED, a new
display type especially suited for small digital camera or mobile phone displays. Design challenges that
came up when negative voltages have to be handled on chip will be discussed, such as
continuous/discontinuous mode transition problems, negative voltage feedback and negative over-voltage
protection. Both devices operate in a fixed frequency PWM mode or alternatively in PFM mode. The single
inductor topology is called inverting buck-boost converter or simply inverter. The proposed converter has
been implemented with a TSMC 0.13-um 2P4M CMOS process, and the chip area is 325 x 300 um2.
KEYWORDS
Buck–boost converter ,DCM / CCM detection, Negative supply voltage.
1. INTRODUCTION
DCDC converters are needed in many battery powered portable products. The battery voltage
varies with load and battery lifetime. Alkaline, Ni-MH,Ni-Cd, Lithium and Lithium-Ion batteries
are widely used today[1]-[2]. Some of them are rechargeable, some are not. Depending on the
application requirements, various DCDC converter concepts are used that regulate the battery
voltage to a fixed desired voltage. Linear voltage regulators have been used for a long time, but
today charge pumps or inductive switch mode power supplies are preferred because of their
higher efficiency [3]-[4]. Also, switch mode power supplies can generate lower or higher
voltages, while linear regulators only can regulate to lower voltage values [5]. Inductive
converters can easily control output currents from almost zero up to several amps. There are
voltage mode and current mode converters, continuous and discontinuous operating modes,
constant frequency, constant on time, constant off time and hysteretic operation, PWM and PFM
control techniques[6]-[7]. This paper is about design challenges for a voltage mode, constant
frequency PWM inverting buck-boost converter intended to generate a negative output voltage
from a positive input voltage.The converter operates with a fixed frequency PWM control
topology and, if power-save mode is enabled, it usesa pulse-skipping mode at light load currents.
It operates with only 500- µA device quiescent current. Independentenable pins allow power up
and power down sequencing for both outputs. The device has an internal currentlimit overvoltage
protection and a thermal shutdown for highest reliability under fault conditions.

2
2. INVERTING BUCK-BOOST CONVERTER TOPOLOGY
A buck converter decreases an input voltage. At least one switch at the input is required to
connect the input voltage to one side of the inductor. Another switch at the same side of the
inductor switches to ground in the off state or alternatively, a diode takes over the decreasing
inductor current. The other side of the inductor is permanently connected to the output.
Acapacitor has to be in place at the input and at the output for stability reasons and to limit huge
voltage drops upon fast load transients. A boost converter increases an inputvoltage. At least one
switch at the output is required to connect one side of the inductor to ground.
Another switch at the same side of the inductor switches to the output in the off state or
alternatively, a diode takes over the decreasing inductor current. The other side of the inductor is
permanently connected to the input. A buck-boost converter basically is a combination of a buck
and a boost converter. There are normally two switches at the input and two switches at the
output. It can either increase or decrease the input voltage. An inverting buck-boost converter has
only one switch at the input and one switch or a diode at the output. But, to be honest, since
integrated circuits usually cannot handle negative voltages, the switch at the output cannot be
used. The diode becomes a necessity. Therefore, sometimes a slightly modified boost converter is
used and another inductor and another capacitor are arranged as shown in fig. 1 to generate a
negative output voltage.
Figure.1. Simple buck-boost converter topology
The inverter discussed in this paper does not require two inductors but uses the simple one
inductor concept shown in fig. 1. Diode D1 just indicates that there is a parasitic pn-junction
associated with the PMOS switch S1 which is the only component in figure that is on chip. The
rest are all external components. D2 is expected to be a schottky diode. Rout represents the load
which could be replaced by a current source Iload. S1 has to be controlled such that the desired
voltage VOUT remains stable under all VIN and Iload conditions. In continuous current mode
CCM operation the inductor current never reaches zero or goes below zero. S1 is turned on and
off with a constant frequency. The duty cycle D determines the output voltage according to
equation (1) or (2).
ܸܱܷܶ = −ܸ‫ܰܫ‬ ×
‫ܦ‬
1 − ‫ܦ‬
(1)
Let D1 be the duty cycle ton/T, for which the switch S1 is on and let D2 be toff/T for which the
inductor current continues to flow through diode D2. In CCM D1+D2=1 or D2=1-D1.

3
ܸܱܷܶ = −ܸ‫ܰܫ‬ ×
‫ܦ‬1
‫ܦ‬2
(2)
In order to maintain VOUT constant, the duty cycle has to be controlled accordingly. The
equation could be solved either for D1 or for D2. Since D1 and D2 are directly dependent on each
other, it does not matter which one of the two are controlled. This means, that if we can set the off
time according to equation (3), VOUT stays constant. Of course, this is true for any value of
VOUT. Therefore, the off time has to be changed immediately from (3) when VOUT changes e.g.
due to load variations. Also, it will never be possible to set the off time accurate enough just by T,
VIN and VOUT. According to (3), the off time would follow a VIN change perfectly and VOUT
would stay unchanged. But, there will always be some regulation that fine tunes the off time.
‫݂݂݋ݐ‬ = ܶ ×
ܸ‫ܰܫ‬
ܸ‫ܰܫ‬ − ܸܱܷܶ
(3)
Figure 2 shows three paths that influence the off time. The “toff control” and “osc” blocks each
use the same type of matched capacitor, which is charged via a current to a reference voltage
proportional to VIN.
The current used in the oscillator is also proportional to VIN in order to keep T constant. The
current in the “toff control” block is made proportional to VIN-VOUT so that toff is
approximately correct even under open loop conditions. Two additional regulation loops control
the off time. One loop is as fast as possible and has to be designed with the constraint to keep the
loop stable. The other loop is compensated externally by a big capacitor and is very slow. It has a
very high DC gain and assures high accuracy.
Figure.2. Off time control loop

4
As can be seen also in figure 2, the error amplifier regulates the FB pin to 0V. This is possible
when the voltage divider that monitors VOUT is connected between VOUT and VREF instead of
VOUT and GND.
As a result, although sensing a negative voltage, no negative voltage value has to be applied to the
chip.
3. DISCONTINUOUS OPERATION ISSUES
Usually, in the application the load changes between no load and very high peaks that can be
covered only by the load capacitor. Current limit circuits protect the chip in overstress or short
circuit cases. Reducing the load would result in negative current flow but due to the presence of
D2, the inductor current becomes zero. Figure 3 shows a CCM example, while figure 4 shows
DCM inductor current and switch pin SW behavior.
Figure.3. Continuous conduction mode CCM
Figure. 4. Discontinuous conduction mode DCM
In contrast to CCM where the off time is independent of the load, the off time in discontinuous
mode no longer follows equation (3). With D1xT being the time for which the switch is on and

5
D2xT being the time for which the diode conducts, D1+D2 is no longer equal to 1. As mentioned
earlier, it is not possible to use a switch instead of a diode for an inverter. A NMOS device cannot
be connected to a negative voltage. The pn-junction to the substrate would clamp the voltage to
about -0.7V. A PMOS device would have to be controlled by negative voltages below VOUT to
turn it on which is not possible, too. Even if NMOS devices could be isolated from substrate,
there is no way to turn it off with normal logic levels. Level shifting to –VOUT is not really a
solution to control big switches. Thus, the diode cannot be replaced by a switch which
automatically forces discontinuous operation at light loads. Now, since discontinuous operation
cannot be avoided, let us derive the equation that is valid for the off time in that mode.
As can be seen in figure 4, (D1+D2) < 1. The net volt-seconds area across the inductor is zero.
VIN × D1 × T + VOUT × D2 × T = 0 (4)
Solving for VOUT results in
VOUT = −VIN ×
D1
D2
(5)
One way to determine D1 or D2 is to consider the power balance between the input and output.
With the average inputcurrent IIN and the average output current IOUT and 100% efficiency
VIN × IIN = VOUT × IOUT (6)
Using equation (5)
IIN = −IOUT ×
D1
D2
=
−VOUT
ROUT
×
D1
D2
=
VIN
ROUT
× ൬
D1
D2
൰
ଶ
(7)
The average input current plus the average output current is the same as the average inductor
current. Let the peak inductor current be ∆IL and the period for which this current flows is
(D1xT+D2xT). This period is the base of a triangle that defines the inductor current. The average
inductor current is obtained as the area of this triangle divided by the clock cycle period.
IIN = ∆I୐ ×
D1 + D2
2
− IOUT (8)
Equating equations (7) and (8) delivers
∆I୐ =
2 × −VOUT
ROUT × D2
(9)
Also, we know
∆I୐ =
D1 × T × VIN
L
=
D1 × VIN
f × L
=
D2 × −VOUT
f × L
(10)
Substituting for ∆IL from equation (10) in (9) we get
D2 × −VOUT
f × L
=
2 × −VOUT
ROUT × D2
(11)

6
Equation (11) can be re-written as
(D2)ଶ
=
2 × f × L
ROUT
(12)
Solving for D2
D2 = ඨ
2 × f × L
ROUT
(13)
The off time, our parameter to be regulated is
toff = T × (1 − D1)(14)
Substituting for D1 from equation (5) and for D2 from equation (13) we get
toff = T −
−VOUT
VIN
× ඨ
2 × T × L
ROUT
(15)
As can be seen easily when we compare equations (3) and (15), the off time behaves very
differently in continuous and discontinuous operation. Equating D2 in continuous mode which
can be derived from equation (3) with the expression for D2 in (13) and solving for ROUT gives
the CCM/DCM transition point value for ROUT.
In discontinuous mode, the error amplifier has more work to do in regulating load changes. As a
result, the fast error amplifier path is not able to regulate out fast load changes whenever CCM
OR DCM regulation is left or entered. It takesa while to bring VOUT back to the desired value
via the slow, high gain error amplifier path. Huge output voltage drops can occur due to sudden
load jumps as can be seen in figure 5 where VOUT has been measured when the output load
suddenly changes from 10mA to 120mA. A drop of 2.5V with VOUT=-12V means about 20%.
Figure 6 shows the VOUT overshoot from 120mA back to 10mA.

7
Figure .5. VOUT voltage drop from -12V to -9.5V upon sudden load change from 10mA to 120mA
Figure.6. VOUT voltage overshoot from -12V to -13V upon sudden load change from 120mA to 10mA
In order to improve transient load behavior, a simple but effective solution has been found. To
keep the regulation loop stable, a large compensation capacitor is required in CCM, but not in
DCM. Therefore, the external big capacitor is disconnected in DCM. However, this is possible
only if the external capacitor starts with the correct voltage immediately when entering CCM
again. Best results have been achieved by making sure that the voltage across the external
capacitor follows the internal much faster voltage signal all the time. Thus, switching back to
CCM always starts with the correct operating point. Switching back to the large capacitor is a
little bit delayed to allow the voltage to come up fast in case of a significant load step. Switches

8
as shown in figure 7 and a special low power output buffer are needed to improve transient load
behavior. The ratio of the external capacitor to the internal capacitor can be 100 or even more.
Figure.7. Transient load response improvement
The transient load response improvement reduces the output voltage drop from 2.5V to 0.6V as
can be seen in figure 8. This means a 5% drop instead of 20%.
Also, the overshoot reduces significantly as can be seen in figure 9.
The transition point from CCM to DCM and vice versa can easily be detected by monitoring the
SW pin. DCM means the inductor current becomes 0. Just before the current becomes 0, the
voltage across the inductor is still VOUT. Since the voltage cannot jump to 0V immediately the
current continues to flow for a moment and moves the voltage from VOUT to VIN before finally
it can drop ringing to 0V. This first voltage jump can be detected easily as shown in figure 10.
A simple PMOS source follower provides the level shifting and due to capacitive coupling the
propagation delay is only a few nanoseconds. An anti-series combination of two diodes protects
the MOS devices from ESD strikes without clamping negative voltages too early.

9
Figure.8. VOUT voltage drop from -12V to -11.4V upon sudden load change from 10mA to 120mA with
transient load response improvement
Figure.9. VOUT voltage overshoot from -12V to -12.4V upon sudden load change from 120mA to 10mA

10
Figure.10.DCM / CCM detection
4. OVER-VOLTAGE PROTECTION
To protect external circuitry from negative voltages far above what can be tolerated, the circuit
shown in figure 11 has been designed for solution where a separate comparator monitors the
voltage feedback pin FB and stops switching whenever the output voltage exceeds the desired
voltage by more than 10%. The additional second protection circuit directly monitors the negative
output voltage. This makes sense for the case when the resistive voltage divider is broken. The
output voltage limit has been set to -20V by choosing the resistor ratios accordingly.
Figure .11. Over-voltage protecti
Figure .11. Over-voltage protection

11
5. FIXED VOLTAGE VERSIONS
Any output voltage from -2V to -15V can be generated by setting the appropriate values for the
external voltage divider. Fixed voltage versions e.g. -5V, -12V and -15V do not need the external
resistors. In contrast to converters that generate a positive output voltage, it is not possible to
program various versions by using EEPROM bits. NMOS devices cannot be used to switch
negative voltages without forward biased pn-junctions or they simply cannot be turned off.
PMOS devices cannot be used to switch negative voltages without having an even more negative
gate voltage applied. There is no practical solution. Therefore we decided to offer bond options
that are easy to realize and do not add any new risks. Since copper bond pads can be placed over
active area, there is no extra area consumption. Figure 12 shows theFixed voltage versions by
bond options
Figure .12. Fixed voltage versions by bond options
6. LAYOUT
The Layout of the proposed inverting buck-boost DCDC converter is shown in Figure. 13. Layout
techniques such as guard rings and careful power and ground routing can minimize the spikes
induced onto the sensitive linear sections.

12
Figure.13. Layout of the proposed inverting buck-boost DCDC converter
7. CONCLUSION
Thedesignofan inverting buck-boost DCDC converter designwith a fixed-frequency PWM control
topology for small digital camera or mobile phone displayshasbeenpresented. The proposed
simple non-inverting buck-boost converter topology can be used successfully to generate negative
supply voltages. Of course, using a diode instead of a switch costs efficiency, but we have
achieved more than 80% in CCM which is 10% more than the competition device.
ACKNOWLEDGEMENTS
This workwassupported by: le Centre National de la Recherche Scientifique et Technique
(CNRST Maroc) under the TIC R&D program.
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